How to Configure I2C Sensors with Arduino

I’ve spent the last year in the ‘uncanny valley’ of the Arduino. That’s the point where you understand the tutorials at Arduino.cc, but still don’t get much from the material on gitHub because trained programmers would never stoop to using the wire.h library when they could just roll their own in native C++ using the avr-g compiler.  The problem with establishing sensor communication at the level of the TWI peripheral inside the AVR is that there are so many fiddling details to keep track of that it quickly overruns the 7±2 things this average human can hold in his head at one time: Computers aren’t the only things that crash after a buffer overflow!  So this post is meant to be a chunking exercise for beginner-intermediate level people like myself who want to get a new sensor working using the standard IDE.  I’ve tried to distill it all down to things that I run into frequently, but there’s still a lot of material here:  So pour yourself a cuppa before diving in...

The great strength of I2C is that you can put so many sensors on the same four wires. But for units with several pre-made modules connected you might have to remove a few smd resistors from the breakouts, or the pull-up on the bus might become too aggressive. Most of the time I just leave them all on, so I can extend the wire length, or crank up the bus clock…

REGISTERS are simply memory locations inside an I2C device. The summary of how many registers there are in a given sensor, and what they control or contain is called a register map. Most of the information on the sensor’s datasheet is about explaining how each register functions, and they can be quite a slog to read through because the information is rarely presented in an intuitive way.

To give you a sense of what I mean by that: take a look at page 14 of the manufacturers datasheet for the ADXL345 accelerometer:

A document only a hardware engineer could love…

Then take a look at the interactive register map for that sensor over at the i2cdevlib site:

Even if you’ve never worked with registers before, jrowberg’s visual grid layout makes it easy to see how the sensor’s memory is divided into sections, which are doing different things.

There are many kinds of registers but for this introduction I am going to group them into three general types: Control, Data and Status registers, and provide brief examples of code that you can use to work with each of them. The functions named with the i2c_ prefix should be generic enough to work with most I2C sensors, but I’ll also be referring to a few specific cases to show how you might need to modify those basic functions.

1) Control Registers

Most sensors change how they operate based on the values stored in control registers. Think of control registers as banks of On/Off switches, which you turn on by setting a bit to 1 and turn off by setting that bit to 0.  I2C chip-based sensors often have a dozen or more operational settings for things like bit-depth, sampling speed, noise reduction, etc., so you usually need to set bits in several different control registers before you can actually take a reading. And sometimes there are “special chip functions” that perform some kind of post processing on those sensor readings that would be hard to replicate on the Arduino. These can add an extra layer of control settings to take care of when you initialize the sensor.

Arduino’s wire library can only transfer 8-bit bytes over the I2C bus, so that’s the smallest amount of information you can write into a register memory location at one time. This can potentially change eight of those control switches simultaneously and, for parameters that are controlled by more than one bit, sometimes it’s actually required that you set them in one register-writing operation.  Most people use byte variables for the sensor’s bus and register memory addresses, but once you’ve figured out the pattern you need to set up in control register switch-bits, it helps to write that information as a long form binary number (eg. 0b00001111) so you can see the on/off states when you read through your code. 

Writing a byte to an sensor’s control register can be done with four basic steps:

Wire.beginTransmission(deviceAddress);  // Attention sensor @ deviceAddress!
Wire.write(registerAddress);   // command byte to target the register location
Wire.write(dataByte);                           // new data to put into that memory register
Wire.endTransmission();

The I2C deviceAddress is set by the manufacturer but some can be modified from their defaults by connecting solder pads on the breakout board.  Since the bus address of a given sensor IC can vary from one module to the next I keep Rob Tillaart’s bus scanner handy to find them, and more importantly to discover when two sensors are fighting with each other by trying to use the same address on the bus.  The registerAddress moves a pointer inside the chip to the memory location you specified. You can think of this pointer as a read/write head and once that pointer is aiming at a specific register, the next byte you send along the wires will over-write the data that was previously stored there.

The startup default values for a given control register are often a string of zeros because all the chip functions being controlled by that register are turned off. Unfortunately this means you’ll find lots of poorly commented code examples out there where people simply write zero into a control register without explaining which of the eight different functions they were aiming for because seven of those were still at their default zero-values anyway.

Reading data from a sensors memory register(s) requires two phases:

Wire.beginTransmission(deviceAddress);    // get the sensors attention 
Wire.write(registerAddress);    // move your memory pointer to registerAddress
Wire.endTransmission();           // completes the ‘move memory pointer’ transaction

Wire.requestFrom(deviceAddress, 2); // send me the data from 2 registers
firstRegisterByte = Wire.read();             // byte from registerAddress
secondRegisterByte = Wire.read();       // byte from registerAddress +1

The first phase tells the I2C slave device which register that we want to read but the wire.library buffers everything behind the scenes and does not actually send anything over the wires until it gets the Wire.endTransmission(); command.  The second phase is the data reading process and you can request as many bytes as you want with the second parameter in Wire.requestFrom. The memory location pointer inside the sensor will automatically increment forward from the initial memory register address for each new byte it sends.

These simple patterns are at the heart of every I2C transaction, and since they are used so frequently, they generally get bundled into functions:


byte i2c_readRegisterByte (uint8_t deviceAddress, uint8_t registerAddress{
byte registerData;
Wire.beginTransmission(deviceAddress);              // set sensor target
Wire.write(registerAddress);                                     // set memory pointer
Wire.endTransmission();
// delete this comment – it was only needed for blog layout.   
Wire.requestFrom( deviceAddress,  1);     // request one byte
resisterData = Wire.read(); 
// you could add more data reads here if you request more than one byte
return registerData;           // the returned byte from this function is the content from registerAddress
}
// delete this comment – it was only needed to maintain blog layout
byte i2c_writeRegisterByte (uint8_t deviceAddress, uint8_t registerAddress, uint8_t newRegisterByte
 {
byte result;
Wire.beginTransmission(deviceAddress);
Wire.write(registerAddress);  
Wire.write(newRegisterByte); 
result = Wire.endTransmission(); // Wire.endTransmission(); returns 0 if write operation was successful
// delete this comment – it was only needed for blog layout.
//delay(5);  // optional:  some sensors need time to write the new data, but most do not. Check Datasheet.
if(result > 0)  
{ Serial.print(F(“FAIL in I2C register write! Error code: “));Serial.println(result); }
// delete this comment – it was only needed for blog layout. 
return result;    // the returned value from this function could be tested as shown above
//it’s a good idea to check the return from Wire.endTransmission() the first time you write to a sensor 
//if the first test is okay (result is 0), then I2C sensor coms are working and you don’t have to do extra tests

//NOTE: copy/pasting code from blogs/web pages is almost guaranteed to give you stray/302 errors because
//of hidden shift-space characters that layout editors insert. Look at the line your compiler identifies as
//faulty, delete all the spaces and/or retype it slowly and carefully ensuring you enter only ASCII characters.


Those two simple functions will let you control the majority of the I2C sensors on the market, provided you can figure out the correct pattern of bits to send from the datasheet. A common strategy for keeping track of complicated multi-bit combinations for control registers is to use #define statements at the beginning of your program, which replace the human readable labels with the actual numbers at compile time.

For example the ADXL345 can range from 3 samples per second to 1600 samples per second, depending on four bits in the ADXL345_BW_RATE register. A set of define statements to represent those bit combinations might look like:

byte ADXL345_Address=0x53;     // the sensors i2c bus address (as a hex number)
byte ADXL345_BW_RATE=0x2c;    // the memory register address
#define ADXL345_BW_1600  0b00001111
#define ADXL345_BW_800    0b00001110
#define ADXL345_BW_400    0b00001101
#define ADXL345_BW_200    0b00001100
#define ADXL345_BW_100    0b00001011
#define ADXL345_BW_50      0b00001010
#define ADXL345_BW_25      0b00001001
#define ADXL345_BW_12      0b00001000
#define ADXL345_BW_6        0b00000111
#define ADXL345_BW_3        0b00000110
etc…. Note that all of these combinations assume normal power mode (bit4=0)

So a command to set the sampling rate to 50 Hz could be written as:

i2c_writeRegisterByte(ADXL345_Address, ADXL345_BW_RATE, ADXL345_BW_50);

 The cool thing about using defines is that they do not use any ram memory like byte variables would. And you can usually find code examples on gitHub where someone has transcribed the entire register address list into a set of defines, which you can simply copy and paste into your own code. This saves you a great deal of time, though there’s always the chance they made a transcription error somewhere. Also note that typical ‘c’ language examples would express those numbers as hex “0x0F” instead of “0b00001111”.

Writing a whole byte to a register is pretty straightforward, but it gets more complicated when you need to change only one of the bit-switches inside a control register. Then the standard approach is to first read out the register’s current settings, do some bit-math on that byte to affect only bit(s) you want to change, and then write that altered byte back into register’s memory location.

But bit-math syntax is one of those “devils in the details” that makes relatively simple code unreadable by beginners. The bit operators you absolutely must be familiar with to understand sensor scripts you find on the web are: the bitwise OR operator [|] , the bitwise AND operator [&], the left shift [<<] and the right shift [>>] operators.  Fortunately there is an excellent explanation of how they work over at the Arduino playground, with a set of bit-math recipes in the quick reference section that let you reach into a byte of data and affect one bit at a time.  Be sure to parenthesize everything when using bitwise operators because the order of operations can be counter-intuitive, and don’t worry if you have to look up the combinations every time because most people forget those details once they have their code working. I know I do. 

Two particularly useful procedures:

x &= ~(1 << n);   // AND inverse (~) forces nth bit of x to be 0. All other bits left alone
x |= (1 << n);       // OR forces nth bit of x to be 1.  All other bits left alone

And these let us add a third function to the standard set which will turn on or turn off one single bit switch in a sensors control register:

byte i2c_setRegisterBit ( uint8_t deviceAddress,  uint8_t registerAddress,  uint8_t bitPosition, bool state )  { 
 byte registerByte, result;
registerByte = i2c_readRegisterByte ( deviceAddress,  registerAddress ); // load the current register byte
// delete this comment – it was only needed to maintain blog layout
if (state) {   // when state = 1
  registerByte |= (1 << bitPosition);   //bitPosition of registerByte now = 1
//or use bitSet(registerByte, bitPosition); 
  }  
else {           // when state = 0
   registerByte &= ~(1 << bitPosition);   // bitPosition now = 0
//or use bitClear(registerByte, bitPosition); 
  }
// now we load that altered byte back into the register we got it from:
result = i2c_writeRegisterByte ( deviceAddress,  registerAddress,  registerByte );
return result;   // result =0 if the byte was successfully written to the register


The ADXL345 accelerometer supports low power modes that use about 1/3 less power than the ‘standard’ operating modes.  The sensor is not turned off, but the bandwidth is reduced somewhat, so the sensor responds more slowly to things like tap inputs.
An example which sets the single bit enabling this low power mode might look like:

i2c_setRegisterBit( ADXL345_ADDRESS,  ADXL345_BW_RATE,  5,  );

Many I2C sensors have power saving features like that which rarely get utilized. Note that bit position numbering starts with 0 and counts from the left OR the right hand side depending on the sensor manufacturer. 

Some devices have control registers that are 16-bits wide. These get treated as a pair of 8-bit bytes that are read from or written to sequentially, but you only have to specify the device & register address once at the beginning of the process because the sensors internal memory pointer gets incremented automatically.

This adds an extra wire.write step to the basic register writing operation:

Wire.beginTransmission(deviceAddress);
Wire.write(registerAddress);
Wire.write(MSB_registerData);    // Send the “upper” or most significant bits
Wire.write(LSB_registerData);     // Send the “lower” or least significant bits
Wire.endTransmission();

The MCP9808 is a common temperature sensor that uses 16-bit control registers.  Setting “bit 8” of the CONFIG register to 1 puts the sensor into shut down mode between readings and setting that bit to 0 starts the sensor up again. (yes, that’s opposite to the usual on/off pattern…)  The 8-bit limitation of the I2C bus forces us to retrieve the register in two separate bytes, so bit 8 of the 16 bits described in the datasheet ends up in the zero bit position of MSB. 

A custom function shutting down the MCP9808 might look like this:  

#define MCP9808_i2cAddress          0x18    // defines in setup are an alternative to using variables
#define MCP9808_REG_CONFIG   0x01    // the compiler swaps the text-name for the # at compile time
// delete this comment – it was only needed to maintain blog layout
void mcp9808shutdown()      //since we used defines, we did not pass any byte variables into the function

 byte MSB, LSB;
 Wire.beginTransmission(MCP9808_i2cAddress);
 Wire.write(MCP9808_REG_CONFIG);
 Wire.endTransmission();
// delete this comment – it was only needed to maintain blog layout
 Wire.requestFrom(MCP9808_i2cAddress, 2); //request the two bytes
 MSB = Wire.read();       // upper 8 bits described in data sheet as 15-8
 LSB = Wire.read();        // lower 8 bits described as 7-0 in the datasheet
// delete this comment – it was only needed to maintain blog layout
 MSB |= (1 << 0); // bitmath forces MSB bit0 (which is ‘bit8’ in the datasheet) to value one
 // using MSB &= ~(1 << 0); here would start the sensor up again by forcing the bit to zero
// delete this comment – it was only needed to maintain blog layout
 Wire.beginTransmission(MCP9808_I2cAddress);  // now write those bytes back into the register
 Wire.write(MCP9808_REG_CONFIG);
 Wire.write(MSB);                          // the one we modified
 Wire.write(LSB);                           // unchanged
 Wire.endTransmission();
}


This ‘position x becomes position y’ translation is common stumbling block for beginners working with multi-byte registers – especially when you add reverse order position numbering into the mix.  But there’s another gotcha with 
control registers that’s even more frustrating if you don’t catch it on your first pass through the datasheet:  Sometimes there are special “write protection” registers that have to be set before you can change any of the other control registers, and these have to be changed back to their “protecting” state before those new settings take effect. You might not get any error messages, but nothing will work the way it should until you get the protection bits disabled and re-enabled in the right sequence. Fortunately less than 20% of the sensors I’ve worked with have this  feature.

Another thing to watch out for are old code examples on the web that appear to be using integer variables to store device and memory register locations, with statements like Wire.send((int)(eepromaddress >> 8));  The I2C wire library only sends bytes/uint8_ts, but people got away with this (int) cast  because it was being corrected behind the scenes by the library, which re-cast the value into a byte at compile time.  The (byte) data type on Arduino is interchangeable with the (uint8_t) variables you see in most C++ coding tutorials. 

2) Data registers

Unlike a control registers bank-of-switches, I think of data output registers as containers holding numbers which just happen to be stored in binary form. Since eight bits can only hold decimal system values from 0 to 255 you usually have to “re-assemble” larger sensor output values from bytes stored in consecutive memory locations. For sensors like the ADXL345 you can concatenate the two 8-bit bytes into one 16-bit integer variable by shifting the MSB left by 8 positions and merging in the LSB with a bitwise OR :

Wire.beginTransmission(deviceAddressByte);  // the pointer setting transaction
Wire.write(registerAddressByte);
Wire.endTransmission();

Wire.requestFrom(deviceAddressByte,2);       // request two bytes
LSB = Wire.read();                                                // byte from registerAddressByte
MSB = Wire.read();                                              // byte from registerAddressByte +1
int combined = (int)MSB;             // MSB now in rightmost 8 bits of combined int
combined = combined<<8;          // shift those bits to the left by 8 positions
combined |= LSB;     // logical OR keeps upper bits intact and fills in rightmost 8 bits

Those steps are usually written in one single line as:

int combined = (((int)MSB) << 8) | LSB;

There are several other ways to combine bytes and some sensors send the MSB first – so you have to check the register map in the datasheet to know the order of the bytes that arrive from the output registers when you request multiple.

Now if you are thinking that looked too easy – you’re right! Most hobby market I2C sensors only have a 12-bit ADC, and since memory is a limited resource there are often status register bits mixed in with the data held in the MSB. Since these bits are not part of the sensor reading, you need to &-mask them away before you combine the MSB & LSB. It gets trickier when the sensor output can be a positive or a negative number because signed and unsigned integers are distinguished from each other by a special “sign” indicator bit, which can accidentally be turned into a “number” bit by bit shifting. (see: ‘sign extension’ in that bit math tutorial )

The temperature data output register in the MCP9808 is a good example of both of these issues:

Bits 15-13 (which become the top 3 bits of the upperByte in the code below) are status indicator flags identifying when high & low temp. alarm thresholds have been crossed. Bit 12 is a sign bit (0 for +ve temperature or 1 or -ve temps). The remaining bits 11-8 (=bits 3-0 of the upperByte) are the most significant 4-bits of the 12-bit integer representing the temperature.

So a sensor-specific approach to reading the temp. from an MCP9808 might look like this:

int TEMP_Raw;
float TEMP_degC; 

// spacer comment for blog layout
Wire.beginTransmission(0x18);    // with mcp9808 bus address written in hex
Wire.write(0x05);                             // and the temperature output register
Wire.endTransmission(); 
Wire.requestFrom(0x18, 2); 
byte UpperByte = Wire.read();          // and sometimes the MSB is called the “highByte” 
byte LowerByte = Wire.read();          // sometimes called the “lowByte” 
// spacer comment for blog layout
UpperByte = UpperByte & 0b00011111;  // Mask away the three flag bits
//easier to read when the mask is written in binary instead of hex
// spacer comment for blog layout
//now we use a mask in a slightly different way to check the value of the sign bit:
if ((UpperByte & 0b00010000) == 0b00010000)  {          // if sign bit =1 then temp < 0°C
UpperByte = UpperByte & 0b00001111;                             // mask away the SIGN bit
TEMP_Raw = (((int)UpperByte) << 8) | LowerByte;    // combine the MSB & LSB
TEMP_Raw-= 256;   // convert to negative value: note suggested datasheet calculation has an error!
 }
else  // temp > 0°C  then the sign bit = 0  – so no need to mask it away
 {
TEMP_Raw= (((int)UpperByte) << 8) | LowerByte;
 }
// spacer comment for blog layout
TEMP_degC =TEMP_Raw*0.0625;


Typically a data output register will continue to hold the last sensor reading until it is refilled with the next one. If your sensor takes a long time to generate this new reading (30-250 ms is typical, while some can take up to a second) and you read the registers before the new data is ready, you can end up loading the previous sensor reading by mistake. That’s where status registers come to the rescue.

3) Status registers

These tell you if if a specified type of event has occurred and I think of these registers as a set of YES/NO answers to eight different questions. The most commonly used status register is data ready [usually labeled DRDY] which sets a bit to 1=true when a new sensor reading is available to be read from the related output registers. Another common status register is one that becomes true if a sensor reading has passed some sort of threshold (like a low temperature alert, or a falling/tilt-angle warning).

A function to check the true=1/false=0 state of a single DRDY bit inside an 8-bit status register might look like this: 

bool i2c_getRegisterBit (uint8_t  deviceAddress, uint8_t  registerAddress, uint8_t  bitPosition) {     
byte registerByte;
registerByte = i2c_readRegisterByte(deviceAddress, registerAddress);
 return ((registerByte >> bitPosition) & 0b00000001);  // or use (bitRead(registerByte, bitPosition))
 }
// delete this comment – it was only needed to maintain blog layout
 //  You could use i2c_getRegisterBit to check the DRDY status bit with a do-while loop
//  and only move on to reading the sensor’s data output registers after the DRDY bit changes to 1
// delete this comment – it was only needed to maintain blog layout
bool dataReady=0;
do {
dataReady= i2c_getRegisterBit(deviceAddress, statusRegAddress, DRDYbitPosition);  
} while ( dataReady==0 );        // processor gets cycled back through this loop until DRDY=1


Holding the processor captive in a status-bit-reading loop is very easy to do, but it is usually avoided unless you are trying to capture a series of sensor readings quickly.  Most status register bits can be mapped to physical alarm output lines on the sensor module, and these can be used to trigger a hardware interrupt pin (D2 & D3) on the Arduino.  This lets you to setup an interrupt service routine (ISR) which grabs that new reading even faster than a bit reading loop. And since hardware interrupts can be used wake a sleeping processor, the interrupt method also lets you put your data logger to sleep until something actually happens. 

The only drawback to the ISR method is that the sequence of register settings you need to follow to create hardware alarms is another multi-step process to add to your sensor initialization code.  The conceptual pattern is usually something like:

  1. Disable the sensor’s global interrupt control bit (if there is one)
  2. Enable the sensors triggering function   (eg:  a max. temperature alert)
  3. Load register(s) with the parameter value for that trigger (eg:  52.5°C)
  4. Turn on the status register that listens to that triggering function
  5. Map that status register bit to a hardware output line
  6. Re-enable the global interrupt control bit

This LSM303 combined accelerometer / magnetometer sensor has two alarm outputs in addition to DRDY. So you could map the Accelerometers DRDY signal to int1, and the Magnetometers DRDY to DRDY.  Just to make life interesting with this sensor, the 3-axis output data  registers are arranged in a different order  on the magnetometer than  they are on the accleerometer. This is typical for multi-sensor chips, which you handle like separate sensors even if they come in the same package – you can even put one to sleep mode while the other one is taking a reading.

Sensors can have many different status monitoring functions, but they usually have only one or two hardware alarm lines.  So the status register -> hardware output mapping (step 5) listed above sometimes involves its own sequence of register settings.  As example, the ADXL345 reads acceleration on three axes, and it has double-tap detection functions for each x,y,z direction. But the Arduino only has two incoming hardware interrupt lines. So generally speaking, you would map all three of those tap-detect status registers to the same outgoing alarm line on the sensor module, and then have the program figure out which axis actually triggered the alarm by reading the status registers later on. High & Low temperature sensor alerts are often mapped in a similar fashion because many breakouts only have one outgoing line: especially if the DRDY status register has been permanently connected to the only other physical alarm line.

A conceptual twist here is that most of the time, the hardware output actually moves the line LOW when the alarm is triggered, even if the status bit it’s mapped from is true=1=high when the actual event occurs. No matter what the status bit->alarm pattern is, any of the four possible interrupt triggers: HIGH, LOW, RISING & FALLING can be used to wake a sleeping 328p processor (though the datasheet states differently).  

Another thing to watch out for on the Arduino side is setting your ISR to respond to HIGH/LOW levels rather than RISING/FALLING edges: Level based interrupts will keep triggering as long as that line is HIGH/LOW. This could cause a sketch to run extremely slowly until the interrupt handler is disabled in your program.  The thing that makes this choice somewhat tricky is that the most common type of sensor failure I see is one where the alarm stays on permanently.  If you set your interrupt to respond to LOW,  and the sensors starts self-triggering your event counters get pushed up to ridiculously large numbers – so it’s very easy to spot that failure in the data, and by the fact that the logger is usually kept awake till the batteries run dry.  If your ISR responds to FALLING, your counts go to almost zero in the same situation, and depending on the phenomenon you are recording it could be very easy to miss that a sensor problem has developed.  Even old analog reed-switched based sensors can suffer from this type of issue, as its not uncommon for something like a wind sensor to stop spinning right where the magnet is holding the reed-switch closed. 

For more information, there’s an excellent guide to interrupt handling over at the Gammon Forum. Probably the most important thing to keep in mind about using interrupts is that by default all interrupts are disabled once you are inside an interrupt subroutine so that the ISR can’t interrupt itself and create an infinite-recursion situation that over-runs the memory.  But the I2C bus relies on interrupts to function, along with timers and other important things.  So don’t try to change a sensor register while inside the ISR,  just set a volatile flag variable and deal with resetting registers later in the main loop.  The general rule of thumb is: “get in & get out ” as fast as possible, and I rarely have a sensor triggered ISR longer than this:

void  INT1pinD3_triggered()  {   INT1_Flag = true;   }

though sometimes I’ll also detachInterrupt(interrupt#) inside the ISR, to make sure it only fires once for things like button de-bouncing. 

Status registers are usually latched, and have to be reset by the I2C master after they are triggered. DRDY registers are cleared by reading information from the data registers they are associated with.  Most other status registers are cleared by reading the register’s memory location, which also turns off the hardware alarm signals that were mapped from them.  This is different from control registers which always have to be explicitly over-written to with new information to change them. If you are waking up a sleeping data logger based on something like a high temperature alert, you usually read the status registers to clear those alarms before enabling interrupts and putting your logger into a power-down state. Threshold based alarms allow interesting things like burst logging.

In Summary:

A good register map, and the four generic functions I’ve described here

  1. i2c_readRegisterByte
  2. i2c_writeRegisterByte
  3. i2c_setRegisterBit
  4. i2c_getRegisterBit

Should be enough to get a typical I2C sensor running, and you can easily tweak those functions to make custom versions for reading 16-bit registers and/or to mask the cruft out of data pulled from mixed registers.

After testing an I2C sensor combination, I pot them in epoxy. Detailed instructions here.

Initializing an I2C sensor is a multi-step process and the correct order of operations is often poorly explained in the data sheet because they are usually written “in reverse”.  Instead of a straightforward list saying “To get a reading from this sensor, do (1),(2),(3),(4), etc.” you find descriptions of the control register bits saying “before you set bit x in this register you must set bit y in this other control register”. When you look up that other control register you find that it too contains a sentence at the end saying “before you set bit y in this control register you must set bit z in this other control register”. So you have to work your way through the document, tracing all those links back until you find the things you were supposed to do first.  Finding the “prime control bit” can be such a time consuming process that it’s not unusual for people who figure out the sequence to wrap it all up into a sensor library so they never have to look at that damn datasheet ever again.

But if you use those libraries, keep in mind that they are probably going to configure your sensor to run at the highest possible bit-depth & data rate, unnecessarily burning away power in applications like data logging which might only need one reading every fifteen minutes.  So the majority of off-the-shelf sensor libraries should be seen as partial solutions, and you don’t really know what else your sensor is capable of until you read through the datasheet yourself.  As an example there are IMU’s out there that will do Euler angle calculations if you simply turn on those functions with the right control register. But libraries for those chips sometimes enable the bare minimum data output functionality, and then do computational handstands to accomplish those gnarly (long) calculations on the Arduino’s modest µC.

In addition there can be useful sensor functions hidden in plain sight, because the datasheet tells you how to turn them on & off, but gives you no clue when to do so. An example here would be humidity sensors like the HTU21D which has an on-chip heating element to help the sensor recover from long periods of condensation, but no status alert that would let you do this automatically. You could just run the heater once a day, but there is also no indication how long the sensor would last if you did that – just some vague references to “functionality diagnosis”. But then some manufacturers (Freescale and Sensirion come to mind…) commit more than just sins-of-omission, breaking away with non-standard I2C bus implementations to lock in customers. The logic there is that if you have to buy the one great sensor that only they make, it’s easier to buy the other four sensors for your device from them as well, rather than juggling low-level protocol conflicts. 

Another challenge when you are working with a new sensor is that Arduino’s C++ environment is not the same as vanilla C in some important ways. So many of the tutorials you find will describe methods that won’t work on an Arduino. Even when the code does compile, there are a number of different “styles” that are functionally identical when they pop out the other side of the compiler, so I’m still trying to wrap my head around the syntax that turns arrays into pointers when they get passed into functionsThat’s why I didn’t mention I2C eeproms in a post about memory registers: almost every multi-byte read/write example out there for EEprom’s uses array/pointer/reference thingies. If you absolutely have to read a series of sensor output registers into an array with a loop, my advice is to just make it a global until you really know what you are doing. And don’t try to store numbers in a char array, because the “temporary promotion” of int8_t’s to 16-bit during some operations can bung up the calculations.

But now it’s time to bring this thing to a close. While I’m still thinking about stuff I wish I’d known earlier, it occurs that a good follow-on to this post would be one about techniques for post-processing sensor data.  There are plenty of useful methods like Paul Badger’s digital smooth, and other code tricks like wrapping those functions in #ifdef #endif statements so those routines only get compiled when a sensor that actually needs them is connected to your logger.

That will have to wait for another day so for now I’ll just sign off with some links. Except for that last ranty bit, I’ve tried to stay out of the I2C handshaking weeds, because when you are up to your neck in bit banging, it’s easy to forget you were trying to measure the water level in a swamp.  But if that’s your thing, there’s some more advanced I2C code examples over at the Gammon Forum, an in depth reference to the Wire library at the Arduino playground , and some troubleshooting tips over at Hackaday

Addendum 2017-11-04

I wonder how many other sensors I could use this with? And if my pin-toggled oversampling method works on the ATtiny, this might provide better resolution than some commercial sensors; though I guess that would depend on how much data I could squeeze into 512 bytes of SRAM…

Somehow I always seem to run into a bunch of related material the day after I post something to this blog: There’s a cool little project over at Quad Me Up using ATtiny85 to turn an analog light sensor into an I2C slave device.  AN4418 from Maxim explains how to use I/O extenders to connect a compact-flash (CF) cards to the I2C interface, which is something I never thought I’d see. And then theres AN10658 from NXP with a method for sending I2C-bus signals over 100m. My own tests with the I2C sensors just hanging off the Arduino only reached about 20m.

Addendum 2017-11-05

Koepel over at the Arduino forum pointed out that the IDE supports some handy macros like bitSet(), bitClear(), and bitRead() that could replace the bit math & masking functions I described here. There’s also word(h , l) to combine two bytes, or highByte() and lowByte() to divide 16-bit variables into 8-bit two pieces. Those were new to me, so I thought I should list them here in case people run into sensor scripts using them.

I’ve also just found out that there are a small number of sensors there that require a ‘false’ modifier to be used at the end of an I2C transaction:  Wire.endTransmission(false);   This is called a repeated start, and the I2C master does not release the bus between writing the register-address and reading data with Wire.requestFrom();   The sensor responds to the I2C address with an acknowledge at the begin of the I2C transaction, and to each databyte that is written to the sensor, so the error code returned by endTransmission can still be used because it is a test if the I2C address was acknowledged by the sensor.

And there was another I2C quirk mentioned at the Gammon Forum:

“You can’t rely on the slave necessarily sending the number of bytes you requested. If you request 10 bytes, Wire.requestFrom() will return 10, even if only 5 have been sent. For the slave to terminate the communication early (ie. after sending less than the requested number of bytes) it would have to be able to raise a “stop condition”. Only the master can do that, as the master controls the I2C clock. Thus, if the slave stops sending before the requested number of bytes have been sent, the pull-up resistors pull SDA high, and the master simply receives one or more 0xFF bytes as the response…It does not help to use Wire.available() because that too will return the number of bytes requested.”

Addendum 2017-11-08

On my page about the DS3231 rtc I describe how to power that I2C chip from a digital pin during bus communication. That trick only works because the chip was designed to gracefully fail over to a backup coin-cell power supply. With other I2C sensors a leakage current might flow into the sensor through the pullup resistors, so you would have to power the bus pullups with the same digital pin to avoid this. And since the internal pullup resistors are enabled by default in the Wire library, you have to disable I2C you could pin power that I2C device.  Also don’t try to de-power a whole module with decoupling capacitors through a digital output pin as that creates big current spikes and really needs proper switching with a PNP transistor of p-channel FET.  99.99% of the time its better to simply find a sensor with a really low sleep current sleep state that you can enter by setting a control register. The best sensors are ones that automatically go into these low current standby-states whenever they detect no traffic on the I2C bus: then you don’t have to do anything.

Another thing I discovered while working with that RTC was that it had a Wire.begin() call hidden in the library, but I was already starting the I2C bus normally during setup. So without knowing it the I2C hardware was being initialized a second time. As the I2C peripheral registers are set to the same value as in the first Wire.begin() call nothing bad happened. However I can see where it might get’s problematic if you call Wire.begin() accidentally because it was buried inside some sensor library while you were running a data transfer,  and the hardware is re-set to an idle state. 

Addendum 2017-11-09

Most of us are familiar with trying out different libraries to drive a sensor, but I’d be remiss if I didn’t mention that there are also some alternatives to Wire library for I2C. The one that gets the best reviews is the I2c Master Library developed by Wayne Truchsess at DSSCircuits. This lib has a faster throughput and a significantly smaller code size: the original Wire library adds about 796 bytes to your sketch when included whereas Wayne’s I2C library only adds 140 bytes. And it has built-in commands that replicate all of the functions I described in this post. For 16 bit registers Wayne points out:

“Technically when sending bytes to a slave device there is no difference between data and an address. In other words let’s say you have a three byte address and three bytes of data. You could use the write(address, regaddress, *data) by making the first byte of your multibyte address equal to regaddress and then combine the rest of the address and data together into *data.”

and that’s equally true with the wire library. The memory savings alone would be worth exploring, but perhaps I2C Master library’s most compelling feature is a ‘TimeOut’ parameter for all bus coms, which could keep your logger from getting stuck in a while-loop if one of your sensors goes AWOL, though I wonder if it still has the 0xFF problem mentioned above, if the sensor sends less than you requested?

Addendum 2017-11-10

I thought using an ATtiny to convert an analog sensor into an I2C device was a neat trick. But it seems that Andreas Spiess has taken the idea to a new level with three HC-SR04’saccessible through on a single AT.  His youTube video #42 with three Ultrasonic Sensors for Arduino walks through the process, with a vocal track that leaves you thinking Werner Herzog has broken into maker videos…

Addendum 2017-11-13

The IDE compiler has an annoying quirk when it runs into Wire.requestFrom in those I2C register routines because the compiler throws up warning messages whenever it feels it has to resolve an ‘ambiguous’ call:  (click to enlarge)

 Turns out that requestFrom has two different implementations, one that can take int arguments, and one that takes uint8_t arguments. If you put in something which has no type like a number (or something you declared with a #define) the compiler has to decide which implementation to use. In the case shown above it chose to use the (int,int) flavor even though device address was specified as uint_8 at the start of the function. 

Anyway, to make those warnings disappear, simply cast the two parameters in Wire.requestFrom to either (uint_8) or (int):

And all those compiler warnings will disappear.

 

 

 

Measuring Electrical Conductivity with Arduino: An Overview

This post is a summary of my background research into electrical conductivity (EC) to serve as a backdrop for my own humble attempts at this interesting measurement challenge. I’m sure there are many other approaches that I’ve yet to discover, and if you know of one please leave a comment so that we can pass that knowledge on to others – Ed.

Obligatory blog-post monkey shot.

Pete & Trish doing profiles with a YSI EXO.

The conduction of current through a water solution is primarily dependent on the concentration of dissolved ionic substances such as salt. Since most fresh water derives from relatively clean rainfall, variations in EC provide a way to track the chemical and hydrological processes the water has been subjected to over time. High amounts of dissolved substances (usually referred to as salinity) can prevent the use of waters for irrigation and drinking, so one could argue that conductivity ranks as the most important inorganic water quality parameter.

A huge number of resources are dedicated to measuring EC and rather than re-hashing all that material, I thought I’d start with links to a few good background reads:

Conductivity, Salinity & Total Dissolved Solids
-discusses the older TDS measurements in part per million (ppm) which makes assumptions about the charge carriers that don’t reflect real world environments.  The conversion factor from EC (which is the thing you actually measure) to TDS changes for different dissolved solids, so instruments from different manufacturers often give you different TDS readings for the same solution, because the companies made different assumptions about what’s in your water.  Because of this confusion, straight EC measurements in siemens have been adopted as the standard by the international scientific community. One siemens is equal to the reciprocal of one ohm (S = 1/Ω)  and is also sometimes also referred to as the mho (℧) in older literature.

Conductivity Theory & Practice
-a white paper that covers basic probe designs, and mentions some non intuitive things like geometry/field effect errors.

Conductivity Sensing at PublicLab.org
-many groups at PublicLab.com have been working on different types of conductivity sensors and their overview page is another excellent introduction to DIY approaches. In fact it’s so good that I will be referring to several of those projects in this post.

Aqueous conductivity is commonly expressed in millisiemens/cm (mS/cm) and natural waters range from 0.05-1.5 mS/cm for freshwater lakes & streams up to about 55mS/cm for sea water. Water up to 3 mS/cm can be consumed, though most drinking/tap water is below 0.8 mS/cm.  Many of the Cave Pearl Project’s installations are in coastal areas where tidally driven haloclines require our instruments to cover that entire “natural waters” range.  Groundwater can vary even more, with measurements being complicated by organic acids and/or significant amounts of dissolved limestone.  Salt water is chemically aggressive and water hydrolyzes above 0.4v, so the probes for high-conductivity environments are usually made of resistant materials such as platinum, titanium, gold-plated nickel or graphite, making them somewhat expensive.

Ways to Measure Conductivity:

There are so many different approaches to measuring EC that it’s taken me a while to digest it all into some working categories.  I expect to build at least one prototype for each of these methods just to see if I can make it work.

Density Based Methods

Refractometers and density based hydrometers are used by aquarium hobbyists. Better quality acoustic doppler flow sensors can also calculate density based on the speed of sound through the water and infer salinity from that. Given then number of acoustic anemometer projects out there, I’m surprised someone has not already adapted the method for underwater applications, though this may be due to the timing limits of the affordable transducers.

Resistance Based methods:

a) Use submerged probes as part of a resistor divider / bridge :
This common approach measures the resistance between two probes using some type of voltage divider. Resistance =1/conductance, which allows you to derive conductivity with your cell K constant since conductivity=(conductance * length)/(area).  AC oscillators are tacked on to reduce electrode polarization, and this forces you to add even more electronics on the output side to convert the signal back to DC for reading. The resistance between the probes changes by several orders of magnitude in environmental waters so different probe surface areas & divider resistors are usually required to cover a significant conductivity range. Above 50% sea water, the resistance between the probes doesn’t change very much, so this method tends to get used more frequently for fresh water environments.

b) Change the pulse frequency of a 555 timer:
You can use resistance between the electrodes as part of an RC relaxation oscillator and then measure the 555’s square wave output frequency to determine the resistance.

This circuit from Thomas Allen’s site provides isolation, uses AC measurement, and the output is a frequency varies from about 42 Hz with the probes in air to > 8000 Hz depending on conductivity.

Circuits and instructions can also be found at at PublicLab.org and there are many good tutorial videos describing 555 based EC sensors on YouTube.  At this point I’ve run into so many projects using this chip that I’d  be willing to bet every environmental sensor I’ve ever heard of could be cobbled together from a few op-amps and a low voltage 555 timer. There are several frequency counting libraries available to help you get started, and if you are ready to sink your teeth into some code, Nick Gammon has produced the some elegant solutions for pulse/frequency timing. Note there are some duty cycle issues

c) Time the discharge of a capacitor through the solution:
Jim Conner’s describes this method in his youTube video at
EC Probes – How they work, and how to build one


A circuit like this might be easy to implement on an Arduino if you can put the internal 1.1v reference onto the comparator that’s also built-in to the 328.  Microcontrollers count time with far better resolution than you get from their ADC’s, but that doesn’t mean there aren’t other issues to deal with. Given that you can try this method with practically no extra circuitry, I will definitely be prototyping a few of these.  Like the 555 based circuits, it will be interesting to see if the method bumps into timing & interrupt handling limits (100 kHz?) when you use it with seawater.

Capacitance based approaches:

You often see capacitance used for liquid level sensors and soil moisture probes, and some of these could be adapted for EC.  To me, the raindrop detection pcb’s you see on eBay have always looked like prime candidates for re-purposing as capacitive sensors.

The circuit described for the Chirp Moisture Sensor uses a fixed resistor and a non-contact probe to create a low pass filter whose cutoff frequency changes with capacitance,  which is affected by the electrolytes in solution. This filters an 1-8MHz square wave and the output voltage is accumulated other side of simple diode peak detector circuit for reading. 

You can also vary 555 timer output frequency by changing the capacitor in the tank circuit, or create more complicated oscillator circuits. No matter which cap-based method you use, the supporting electronics have to be located near the sensor – because just about any length of wire will add enough stray capacitance to throw off your measurements. Another thing to keep in mind is that common ceramic capacitors have some of the worst thermal coefficients and aging effects imaginable. Plastic film capacitors using Polyphenylene Sulfide (PPS ±1.5%) or Polypropylene (CBB or PP ±2.5%) are much better for sensor circuits like this, and having a digital capacitance meter on hand is probably a good idea too.

DIY capacitor plate sensors are going to be small values (picofarads ?) and the resulting RC time constants make these methods more suitable for fresh waters, as this leverages the relatively large solution resistance to give you more accurate interval timing.

Potentiometric (4 electrode) Methods

Four-electrode cells uses two “driver” pins to place an electric field across two other “reading” pins that lie between them: 

This paper describes a DIY 4-probe sensor that was used for soil moisture sensing, and you will find quite a few articles using potentiometric methods over at IEEE and Sensors

Nokia/Apple audio jacks came to mind as soon as I saw this diagram, and they might be available with gold plating.  4-electrode methods often measure the voltage between the read pins, which is divided by the exciter pin current to determine the solution’s impedance = 1/conductance.  To obtain the conductivity, the conductance is multiplied by the cell constant of the inner poles. Tracking the pin current lets you compensate for fouling on the plates, and the method can cover a wide range of conductivity. Like inductive methods, this approach tends to work better as the concentration increases. 

Inductive Methods

The conductivity measurement is made by passing an AC current through a toroidal drive coil, which induces a current in the solution. This induced solution current, in turn, induces a current in a second coil, called the pick-up toroid. The amount of current induced in the pick-up toroid is proportional to the solution conductivity. You get industrial grade performance out of this non-contact method in many different types of solutions, but you also need industrial amounts of power the drive the sender coil so it’s hard to implement on the kind of power constraints you see on stand-alone data loggers.  Inductive sensors require a 3 inch radius from any other surface (bio-fouling?) and you see this pretty clearly in the ‘donut on a stick’ sensor heads. It occurs to me that you see very similar components in a wireless charging system, but there’s a lot of devils hiding in those details – like shielding, etc.  It might be possible to press one of the production line proximity sensor chips into service for a low power solution, or simply try measuring changes in inductance due to the presence of salt water.

Off-the-shelf Solutions for Arduino:   (using 2-Electrode Resistance Methods)

TransparentSinglePixl
Atlas Scientific Conductivity Kit
A complete solution including calibration solutions, a range of probes and code libraries. All parts also sold separately: interface boards are ~$35 & EC probes come in around $120 each but they are durable enough for continuous long term submersion.  I2C data transfer is supported, so resolution is not limited to the Arduino’s ADC.  Whitebox labs Tentacle Shields ($35-$110) provide up to four galvanically isolated channels for full hydroponic rigs. Stand alone BNC carriers for $10.
$200
Gravity: EC Sensor Kit for Arduino (K=1)
Another complete K=1 kit solution, but the probes are not robust enough for long term submersion so several people replace the stock probe with the 208DH which is available on eBay for $35. Arduino ADC reads voltage.  The KnowFlow project uses the full set of DFrobot boards.
$70
Vernier CON-BTA EC probe
This 5v K=1 probe covers 0-20,000 μS/cm  in the high range, and the analog voltage output is read by the Arduino ADC. You need an inexpensive adapter board for the BTC connector, and they provide a basic library. One key feature is built in hardware temperature compensation with a 10k thermistor in the probe head. My tests show this reduces the usual 2%/° C reading variation down to about 0.5%/° C, so you still need to do your own calibration to get high accuracy. Like Atlas Scientific, Vernier has many other interesting sensors that are Arduino compatible.
$115
EC/pH Transmitters
This company offers a range of physically bulky turn-key solutions, with the $70 entry level unit claiming 0-5000 μS/cm (fresh waters) and continuous monitoring. Arduino ADC reads voltage. ~$200 units support PH with isolation.
$70-250
Sparky’s widgets MiniEC
An indie who makes several other useful sensor breakout boards, including PH. You have to build or locate your own probes, though they use a standard BNC connector like most EC probes.  Arduino ADC reads voltage output.
$24
EC-Salinity Probe Interface by Ufire
Designed around an ATtiny configured as an I2C slave, probably using the cap-discharge method.
$14.50
Hanna HI 73311 (K=1) Replacement probes
In the past we’ve used used these epoxy&graphite probes from Hanna DIST5 (HI 98311) and DIST6 (HI 98312) testers, which connect to a standard male audio jack.  You can also re-purpose one of the Vernier ABS/graphite probes if you get a used one cheap on eBay, and the Vernier probes have a 10k NTC thermistor built in, which you can read with a divider.
$55
Comercial Standard Solutions
For fieldwork, it’s often easier to transport the dry packets, and mix them on location.  Atlas sells calibration sets, but at the twice the cost of standards when you buy them in larger volumes.  You can find recipes for homemade calibration solutions at Reefnet Central and PublicLab. For a classroom situation, it’s much cheaper to mix secondary “lab standards” in larger quantities, and then test the resulting solutions with a commercial probe that’s been calibrated against commercial solutions. 5.566g of dry NaCl in 1 litre of distilled water will create an ~10,000 μS/cm solution, which you can dilute down for lower concentration standards.
$14/500ml

This photo from Bitnitting gives you a sense of the physical space needed for the Atlas breakouts and a ‘mini’ form factor Arduino.

Hydroponics hobbyists have putting these kits to good use over the years with notable examples like the long running forum thread on Billies Hydroponic Controller, and the well documented adventures over at the Bitnitting Blog.  The people at OpenCTD and other academic projects have put the Atlas boards into real world deployments.

But to me these commercial solutions still leave you stuck with those expensive electrodes which sometimes cost more money than you would pay for a used 4-pole device. More annoying is the fact that those cell constants do not line up with my goal of measuring the entire “fresh” to “marine” range with one sensor, thought if I could extend it a bit the K=10 probe comes close.  This is illustrated by the following graph from Andy Connelly’s Blog which is worth digging through as he has posted lots of other interesting material on calibration, reproducibility, signal detection, etc. 

 

Of course the cell constant changes as your probes get older and dirtier, so you have to re-calibrate them with standard solutions just about every time you want to take a new reading. I’m pretty sure I will end up making my own probes, probably out of Nichrome 80 wire as the vaping fad has made it common on eBay. Some have had good EC results with gold plated PCB traces. Feedback on the Arduino.cc forum suggests that Platinum-Rhodium Thermocouple Wire is another good option.  I’ve also been wondering about Ag/AgCl which is highly resistant to seawater and is commonly used for non-polarizing electrodes in medical/bio applications. (EKG electrodes?) It might also be a good idea to cobble together a DIY magnetic stirrer, based a PC fan and an old hard drive magnet

DIY 2-Probe EC Circuits

The easiest circuits to build yourself are the 555 timer oscillators, but there are plenty of quad-opamp solutions out there for people comfortable with a breadboard. The oldest example I’ve seen is this one by M. Ahmon from the Sept 1977 issue of Electronics magazine which uses the resistance of the solution to modify opamp output:

This circuit uses the first stage of the quad opamp in a Wien-bridge oscillator, reducing errors caused by electrolysis with a 1-kHz signal that gets attenuated by the solution’s resistance before it reaches the driving amplifier A2.  Pot P1 controls oscillator amplitude, and P2 adjusts gain of A2.  A3-A4 form a precision rectifier giving output voltage equal to absolute value of input voltage. This one chip solution seems to have been the basis for many of the current EC projects on the web, including these two exceptionally well documented examples:

Octavia’s EC/TDS/PPM Meter On Limited Budget

Daniel Kramnik’s Digital Salinometer Project

Similar circuits can be found on the breakout modules from Sparky’s Widgets and DFrobot . Using the solution’s resistance in the feedback divider controlling an op-amp is a neat idea, but having only one opamp there imposes hard limits the range you can measure with a given K value probe. There is a more advanced multi-opamp approach over at pulsar.be that can step over several decades.

On more recent EC projects I’m seeing single supply RRIO opamps for the oscillator & gain stages, which are easier to integrate with battery operated Arduino’s. (though any dual supply opamp can be used as a single supply in a pinch; since voltage is relative the opamp doesn’t know whether V- is a negative voltage or ground) To keep using an AC signal, this requires a virtual GND at 1/2 VCC, but the integration also gives you the option of getting rid of the oscillator entirely, since you can use PWM output as your source.

This is beautifully illustrated by the circuit from bhickman’s Conductivity & Temperature Meter over at PublicLab:

Ranging is accomplished with the (red) bank of R1 resistors, and (yellow)R2’s 5/6 can be substituted in for the probe (R8) with those known resistances can be used to track drift. The AC–DC converter stage is built with precision peak detectors. I think this is the best voltage divider approach I’ve seen to date.  To simplify things a bit, you might replace that output stage with an RMS-DC converter; though I’ve not seen any breakouts for those, and I hate working with raw SMD parts.

Sources of Error: 

Even with a clever circuit like the one above you still need to address things like temperature compensation before you get an accurate, repeatable, and stable device. Electrical conductivity measurements are typically referenced to 25 °C using standard temperature compensation factors (α). The conductivity of natural waters exhibit strongly nonlinear temperature behavior, though in practice linear correction factors are most frequently used.  NaCl-based solutions typically have a temperature coefficient (α) of 0.02-0.0214 (~ 2% change/degree C). So to convert your “ambient” conductivity measurement into 25°C “specific” conductivity, the simple linear conversion is:

EC25=ECambient /[ 1 + α (tambient – 25) ], α= 0.02

Field effect errors are significant, causing read errors if bare 2-pole electrode get within 2-5 cm of the solution container: which will completely mess up your calibration and cell constant determination. This is one reason that virtually every EC probe is encased inside a plastic shroud of some sort. That causes field effect errors too, but at least its the same error every the time, rather than one that varies depending on how far you are from the edge of the beaker. Four probe methods also require a fixed volume of solution between the driver electrodes, so the shroud provides that.

Grazing through the hydroponics forums shows plenty of people struggling with cross-sensor interference.  Most notably when conductivity probes affects the accuracy of a PH probe in the same tank.  Any time two devices are immersed in the same environment differences between them can generate ground loop voltages and induce currents which degrade the readings and exacerbate corrosion.  Sometimes you can address these issues with optical or I2C isolators. One helpful contributor at Arduino.cc suggests:

pH electrodes are very high impedance devices and the cabling and connectors are all important – even flexing a decent cable will distort the readings…. Ground loops are the enemy of pH and any other specific ion electrode. I used them a lot in difficult situations and the most trouble-free solution is always to put a buffer op amp (FET type) as close to the electrode as possible – some commercial electrodes come already equipped. Find a decent op amp like the old MAX406, high impedance techniques like PTFE insulators or simply keep the input pin off the board. Modern FET’s take single-sided supplies and run at better than 2-microamps – a 3.6-V lithium cell will give you in excess of 5-year’s trouble and ground loop -free operation. Once you have buffered the signal, you can use any cable you like. As a bonus, you can convert a pH electrode into an ammonia electrode by separating the water from the electrode with PTFE tape as used by plumbers.”

Well, I think that covers most of the stuff I had in my notes, and hopefully gathering it all here saves someone else from burning away that time. I have been experimenting with conductivity quite a bit lately, and I think I might have  come up with an analog approach that will allow people to play with conductivity on shoestring budgets. I just have a little more calibration testing to do before I let that one out of the bag  🙂

Arduino Data Logger: 2017 Build Update

This base configuration sleeps at 0.25mA or less depending on your parts, so with a 4xAA battery pack it should run for a year.

If you need a logger with a rugged waterproof housing, it’s still hard to beat the crimped-jumpers build released in 2016. However sometimes I just want a quick bare-bones unit for bookshelf test runs while I shake down a new sensor. I can whip up a breadboard combo in about twenty minutes, but they stop working if I bump one of the wires loose. I’ve lost SD cards from this half way through a long term test, and I’ve also run into issues with noise & resistance from those tiny breadboard contacts.

To address this I’ve come up with a new configuration that uses a screw-terminal expansion shield originally intended for the Nano.  This requires a modest bit of soldering, and after some practice, between 1-1.5 hours to finish depending on how many “extras” you embed into the basic three component core. In return for that time you get all the pins broken out, making this approach almost as flexible as a breadboard, and much more physically robust. Pop them into some pre-made boxes and these little guys qualify as deploy-able for relatively stable environments.

PARTS & MATERIALS

Bill of Materials: $8.40
Pro Mini Style clone 3.3v 8mHz $1.85
Nano V1.O Screw Terminal Expansion Board
Note: To save time, you can spend an extra 60¢ for pre-assembled boards by Deek Robot, Keyes, & Gravitech.  Note that bad vendors show photos of the pre-assembled boards in their listing, but then ship you the no-name assemble-it-yourself part kit. That kind of bait-n-switch tactic is very common with dodgy eBay suppliers.
$1.05
DS3231 IIC RTC with 4K AT24C32 EEprom (zs-042)
Some ship with CR2032 batteries which will pop if you don’t disable the charging circuit!
$1.25
SPI Mini SD card Module for Arduino AVR
Be sure to buy the ones with four ‘separate’ pull-up resistors for easy removal.
$0.50
4xAA 6V Switched Battery Holder
The logger works with battery packs holding 3 to 8 AA batteries (with the default MIC5205 regulator)
$0.75
CR2032 lithium battery  $0.40
Sandisk Brand Micro SD card 128mb-1gb 
Older Sandisk cards have lower sleep currents. Test used cards well  before putting them in service.
$2.00
Common Cathode Bright RGB LED 5mm 
( & 30kΩ limit resistor)  A brighter bulb lets you use a larger limit resistor for the same light output.
$0.05
Double Sided Tape,  2x 10MΩ resistors, 28awg silicone wireheader pins, etc… $0.50
Donation to Arduino.cc
If you don’t use a ‘real’ Promini from Sparkfun to build your logger, you should at least consider sending a buck or two back to the mothership to keep the open source hardware movement going…so more cool stuff like this can happen!
$1.00
Comment:   You might need one of these to get started:                            (not included in the total above)
CP2102 USB-UART Bridge module
These work with Macs & Windows machines after you install the drivers. Or try the FTDI version.   ***Be sure to set the 3.3v jumpers on the module before using it!***
$2.20

COMPONENT PREPARATION

The Main Board:

In this build the six serial UART I/O pins must have 90 degree angled headers to make more room for the RTC board which will sit on top of the main board later.  Solder those header pins onto your Arduino board, and test it with your UART adapter. Generally speaking, about 10% of the cheap modules I buy from eBay are flakey in some way, and it’s quite annoying to discover that after you’ve assembled a logger. Once you know the board is working, remove the power and pin13 LED resistors.  These limit resistors tend to move around from one manufacturer to the next, so you might have to hunting for them on your particular board.  You also need to remove the RESET switch from the board, or that button will be compressed when you put the SD card adapter into place:

{Click any images to see larger versions.}

pwrledpromini

Solder the side rows of straight header pins so that they project from the bottom of the board.  I usually skip the two reset pins, so that I can re-purpose those screw terminals later as GND and Vcc (photo 3 below) but if your application needs reset functionality then  solder those headers as normal.  Add wires to the top of the board for the A4 (SDA white) & A5 (SCL yellow) lines of the I2C interface.  Add wires to the A6 & A7 vias so that they project from the bottom of the board.

Once all the pins are in place clean any flux residue from the board with 90% isopropyl alcohol and a cotton swab. The final step for the main board preparation is to trim the pin header solder points on the TOP of the board flush with the surface: D4-D9, D10-D13, and A0-A1.  Then affix some double sided tape  in place over those trimmed pins, which will mate with the bottom surface of the SD adapter.

The RTC Module:

The simplest modification to these DS3231 RTC boards is to remove the charging circuit resistor and power LED limit resistor from the circuit board (indicated with the red squares in the first picture).  LIR2032 rechargeable batteries are nominally 3.6v, and will not charge with this module connected to a 3.3v Arduino. Replacing that with a CR2032 will backup the RTC for many years of operation 

rtc1

Add two layers of double sided foam tape, so that the thickness matches the top surface of the DS3231, and the inside edge aligns with that side of the chip. These two surfaces will mate with tape on the SD adapter board.

Since the RTC board already has 4.7k pullups on the SDA (data) and SCL (clock) lines, you will not need to add them to your I2C sensors.  This board also has a 4.7k pullup on the SQW alarm line.  We will be connecting SDA, SCL, GND and VCC wires to the small cascade port on the module.

The SD Card Adapter:

This SD card adapter comes with small surface mount pullup resistors on the MOSI, MISO & SCK (clock) lines (removed from the dashed red line area photo 2 below).  The Arduino SDfat library uses SPI mode 0 communication, which sets the SCK line low when the logger is sleeping. This would cause a constant drain (~0.33mA) through the 10K SCK pullup on the module if we did not remove it.  I prefer to pull MOSI & MISO high using the internal pullups on the 328P processor, so those physical resistors on the breakout board can also be removed. But be careful to leave the top-most resistor of the four in place to pull up the DAT1 & DAT2 lines.  This keeps those unused pins on the μSD card from floating when the cards are accessed in SPI mode.

sd1 Only remove the bottom three pullup resistors. keep the top one

Add jumper wires to each of the headers pins on the bottom of the SD adapter and trim those solder joints till they have a relatively low profile . Then cut away the vertical header pins from the top of the board. Place a strip of double sided tape on the bottom of the SD card module opposite the soldered wires. This strip acts as a spacer to level the SD board when it is placed in contact with on the double sided tape on the mini style Arduino board.

The SPI connections:
RED:           3.3v regulated
Grey:          Cable select (to D10)
Orange:     MOSI   (to D11)
Brown:      SClocK (to D13)
Purple:      MISO   (to D12)
BLACK:     Ground

The Screw Terminal board:

These screw terminal boards are designed for use with Arduino Nano boards, but if you orient the two correctly when you connect them, labels on one side of the shield will be in alignment with promini pins:

 

Drilling a pass-through hole lets you bring the jumpers down to those unused pins, and to make other connections to solder points on the underside of the shield without blocking the M3 mounting holes.

ASSEMBLING THE LOGGER PLATFORM

Attach SD adapter to the Pro mini:

The first step is to attach the SD adapter to the board, but this must be done with a slight overhang, so that at least the red Vcc wire on the SD adapter extends beyond the top surface of the pro-mini board.  It’s OK to leave more overhang than I’ve shown here, but if you leave less, the wires on the RTC cascade port might interfere with access to the serial I/O pins.

Place a strip of double sided tape across the SD adapter board as shown, taking care not to cover the hole showing the card lock spring.  When that tape is in place, bring the ground and Vcc lines from the SD board forward and make a gap in wire insulation so you can splice-solder them to the GND & Vcc pins on top of the pro mini board. This procedure simultaneously connects the rails to the SD adapter, and brings power to the I2C cascade port on the RTC module.

Connect the RTC board:

I recommend that you take a bit of time holding the RTC board in place over the SD & mini combination while the protective covering is still on the tape, so that you get a feel for the alignment before you actually try to stick these parts together. With the cascade port oriented towards the Arduino’s serial I/O pins, the topography of the SD adapter fits snugly into place against the DS3231 chip on the RTC module.

After you stick the pieces together, trim and solder the I2C bus wires to the RTC’s cascade port. Note that it is possible to unstick the parts afterwards by gently levering them apart with a screw driver, but be careful you don’t rip the metal shield off of the SD card adapter in the process.

Attach everything to the Screw Terminal Shield:

If you’ve gotten this far, then you can now relax, because all the tricky stuff is done.  Trim and tin the four SD lines and bring the down to the D10-13 SPI screw terminals just below. Note that D12(MISO)  & D13(SCLK) lines must crossover.  Bend the pins on the RTC board downward and solder jumpers onto all but the 32K output line.

Pass the SQW alarm line (in blue) through the hole and solder it to the D2 pin projecting from the underside of the terminal adapter board.  If you left out the reset pins when initially soldering the headers, bridge those unconnected terminal screws to the adjacent Vcc & GND lines.  Then patch A6/A7, and the four I2C lines from the RTC board to the unused pins at the end of the screw-terminal shield.  I generally run these loggers on 3xAA battery packs with a 2x10M ohm voltage divider providing 1/2 of that battery voltage to A0.  So the last step is to add that voltage divider, along with some extra tape to serve as foot pads.

The battery voltage calculation for a divider with equal value resistors is:   float batteryVoltage = float((analogRead(A0)/ 511.5)*3.3);  But the MIC5205 regulator found on most promini style boards will accept anything between 3.4 to 12v input, so you will need size your resistors to convert the peak battery pack voltage into something below the 3.3v aref limit. To cover that whole range, you’d need a pair that puts 1/4 of the battery voltage on A0, and a R1(high side) = 3*R2(low side) combination would do that, changing the 511.5 constant in the equation above to 255.75    With 5205’s dropout potentially rising to 300mV during 200mA SD writes, I usually shut down the loggers when the main battery falls below 3.75 volts. With Meg-ohm size resistors, I leave that divider connected all the time, but there is a wonderful self-disconnecting voltage divider idea over at JeeLabs for those who want to use smaller resistance dividers.

As we removed the pin13 LED back at the start,  solder a limit resistor onto the ground of a common cathode RGB and connect that to one of the ground connections, with the other legs going to D4R-D5G-D6B.  I usually add a few labels to keep track of the extra terminal connections, and any re-allocated any pins for a specific build. Unfortunately black sharpie marker doesn’t stick to those green terminal shrouds very well. 

In this example I’ve re-allocated the screw terminals that would normally have been connected to the two reset pins, but you could use under-board wires to re-assign any of the terminals in a similar fashion. For example, if your application will not be using the RX/TX pair, those could be turned into extra Ground or Vcc points. I’ve never understood why the pro-mini design breaks out reset twice but leaves the Aref pin hidden, so adding a wire to the little aref stabilizing cap would let you fix that issue.

Your Logger is ready to go!

As this is simply a different physical arrangement of the same core components, you can follow the logger testing procedures described at the end of  the 2016 build post , which also provides links to a basic data logger script to help you get started on your project.  For the build described above, the pro-mini’s MIC5205 regulator delivers sleep currents less than 0.25mA (Promini~0.05mA,  sleeping SDcard~0.08mA & RTC~0.09mA)
That should should reach a year of operation on 4xAA’s. 

While it took me a day to get the first one of these sorted, the second one took less than three hours, and the third took less than 2 hours. I lost count after that, and now these things seem to be multiplying like tribbles.   If you need unobstructed access to the SPI bus, you can move the SD lines to under-shield solder connections as we did for the I2C bus.  This makes the logger a little prettier, but since I’m usually making these in a hurry, I often leave those wires on the surface.

The photos in this series were made with Adafruits 26AWG silicone wire, but if you are adding more bottom-side connections, switch to smaller diameter 28AWG wire, or make the pass-through hole a bit larger to accommodate the extra lines. Switching the  90° I/O header pins to the bottom of the promini board gives you more room for the RTC wiring.

You can make the component stack more rigid by adding a few strategically placed beads of epoxy putty.  In fact you could hold the whole thing together that way, so long as you take care not to bridge any contacts – especially where the DS3231 legs come near the metal top of the SD adapter. Also keep in mind that the putty sets rock hard in about five minutes, so if you make a mistake with that assembly method then you’ve bricked the unit… literally.

Addendum 2017-06-21:

I’ve been on a steep learning curve since the beginning of this project, and you don’t have to dig very far to find stuff on this blog that seemed like a good idea at the time, but later turned out to be completely wrong.  I should write some sort of disclaimer,  but instead I’ll pass along a recent forum comment that summarizes the kind of criticism we’ve been getting lately:

“In the old days, an embedded enthusiast would have designed the thing (and think AVR) from the outset to meet objectives / specs, not struggle with integrating the various modules and meeting very-so-so sleep currents (while thinking Arduino). Surely, this is a textbook example of how not to do embedded engineering if you are doing it for a salary.”

It’s good to have someone rattle your cage once and a while, and I’ll admit they have a valid point( In addition to the fact that I’m not an engineer, and I don’t get paid…)  People complain like that about the pitfalls of using modules & libraries all the time, but the thing I like about the Arduino platform is that you don’t have to know everything before you can do anything. I’m just figuring it out as I go along.

Still, an affront like that demands some kind of response.  So to defend the honor of my fellow Arduino Kool-Aid drinkers, let’s look at how you might tweak those modules to improve this loggers sleep current performance:

1) Pin Power the RTC:

These DS3231 boards don’t get a lot of love because they have about the worst battery charging circuit ever devised, and an equally useless LED power indicator. But for less than a buck delivered to my door, these boards are considerably cheaper than the components they carry: so I’m going to look under that rock and see what I find.  That charger can be disabled with a simple flick of the soldering iron, and at this point we have years of successful run time using a non rechargeable CR2032 as backup.

More interesting is the fact that on a 3.3v system, you can leave the charger in place, cut the Vbat line, and patch in a 1N5819 Shottky.  After the CR2032 burns down the two circuits should balance out, and the main battery takes over supplying the 3µA timekeeping current.  CR2032s are rated for reverse currents on the scale of Shottky leakage, but I’m sure if you ask an engineer they would tell you this is a bit dodgy.  Since I don’t know any better, I’m just going to do it anyway and see what happens…

The final step is to lift the Vcc leg on the IC and jumper it directly to a digital pin to provide power during I2C communications. During sleep this power-pin is driven low: forcing the RTC into backup powered timekeeping mode and it can still provide wake-up alarms to your logger in this state.  These mods cut your loggers sleep current by about 0.1 mA

2) Buffer your data before saving:

Those DS3231 RTC modules also have a 4K EEprom on the board, and that lets me save data in 32byte page-writes with reasonably simple code. While the I2C bus is dead dog slow by embedded system standards, you can hang oodles of things off those wires without worrying about cable select lines, or some gummy protocol weirdness.  For an extra buck, you can add 32K more memory without any significant changes to your Arduino script. That usually buffers about a week’s worth of data before I need to save to the SD, even though I’m still making the unforgivable programming sin of storing everything in ASCII string variables

Small red-board versions of the AT24C256 tuck nicely into the 12mm gap between the headers, but you could just as easily put an I2C sensor into that space. If you get boards with the address pins broken out (the one above doesn’t), you can connect up to four of these eeproms to the same logger. A side benefit is that the 32K eeproms are rated to 400kHz, while the 4k’s are only 100kHz, so the upgrade also lets you accelerate the I2C bus clock, since the DS3231 is also rated for 400kHz.  Even larger eeproms are available in the code compatible AT series, but I’m not sure if the wire library supports the 64byte page writes they typically use. If I had the chops, the path to an IC-only logger is obvious. Paul Paul Stoffregen’s SerialFlash library is another interesting option, as it allows one to read/write files in a similar way to an SD card.

3) Cut power to the SD card:

The clones I’m using have a tap at the back that is conveniently located for ground side switching on the SD cards. This lets me tuck a 2N2222A under the board with that extra eeprom.  Cards hit the reg pretty hard when they initialize causing significant vdrops, and if you find that your unit is not saving properly with this technique it’s probably because those low-voltage transients are restarting your logger.  I usually add caps to provide at least 30μF on the rails to handle those spikes, and I may bump that even higher for cold climate deployments since cheap ceramic caps have terrible temperature constants.

Code and information about this technique are described in some detail on the
SD power post This is a relatively high risk strategy, but it can cut your sleep current by another 0.1mA. Note that while the BJT works OK, on more recent builds I’ve switched to the Supertex TN0702 mosfet for SD power control.

4) Replace the voltage regulator:

The MIC5205 on those pro-mini clones is not very efficient at low power (~10-20% efficient), so replacing that with an MCP1702-3302E/TO can cut your remaining sleep current in half.  The 10uF caps from the original regulator are still in place on the board, so it’s been working fine with the 1700 just hanging off one side. Also keep an eye on the dropout voltage, which on the MCP1700 series can rise as high as 600mv if you push them to their 250mA maximum: requiring a fairly high 3.9v as your input cutoff. If your not ready for soldering in close quarters, you can always dead-bug the reg & voltage divider onto the battery connector like I did for the 2016 builds.

And the result?

This logger is drawing less than 0.02mA sleep current with a MS5803 sensor in tow. That’s 5x more than you’d see from a raw 328p, but not bad considering that we built a fully functional data logger out of 99¢ eBay modules. (With the default 5205 reg in place, the logger would draw ~0.06mA)

These modifications to the basic build plan probably violate some important electrical engineering rules, and I can almost guarantee that nothing will work properly the first time you try it.  But don’t let the fact that you might destroy a few cheap components along the way prevent you from just going for it.  Although it might be best if you don’t show your project to any engineer friends at the beginning… unless they’re working on a new textbook 🙂

Addendum 2017-10-02:

If you are careful about placement of the batteries, you can fit this new screw-terminal design onto the abs knockout plugs that I’ve been using as mounting platforms. This means that you can still fit the logger into the inexpensive 4″ housings
that I outlined for earlier builds with room under the platform for a second battery bank if needed.  Given how often makers need to put a shell around their projects, I’m surprised that no one has taken the old B-Squares idea into three dimensions to create a re-configurable snap-together housing system.

In addition to the I2C bus, I’ve started breaking out A1-A3, rather than digital lines, since those A ports can do double duty as either analog or digital I/O with some code settings. With screw-terminals on all lines, I only break those out to connectors for quick sensor swaps in the field.

One thing to keep in mind for any project built from eBay parts is that most of those boards use cheap Y5V capacitors; which have terrible temp-coefficients compared to X7R/NPO’s.  So you need to test your project extensively if you want it to operate over a wide temperature range. My home freezer tests to date have been running ok, but I’m not relying on the Arduino’s oscillator/clock for anything that is timing-critical.  I do expect to see things like bus timing drift out of spec eventually.  For loggers built with I2C sensors, stick with 100kHz for your first few builds, then things “just work every time”, 75% of the time. The 1.1v internal bandgap also changes significantly with temperature, so if you use it as Aref, expect to see the readings go up, as the temperature goes down.

 

Switching off SD cards for Low Power Data Logging

This composite graph compares logger current during identical sd.begin(), File Open & File Close sequences with two different Sandisk brand SD cards on the same datalogger. The 256MB card used less than 3mAs, while the 2GB card burned more than twice as much power during this initialization. Older low cost/size cards often perform better in SPI mode, which is simply an after-thought for high end cards, because it’s required by the SD spec.

The tweaks I discuss in the  power optimization post bring sleep current on a typical build into the 0.1-0.12mA range; leaving the sleeping μSD cards  as the largest remaining power consumer on the Cave Pearl loggers.   And those cards have been a burr under my saddle for quite some time now, as they are probably responsible for most (if not all..) of the power consumption irregularities that were showing up in some of the battery voltage logs.

I already knew the various brands of SD cards could have dramatically different sleep current, but a comment in the Arduino.cc forum by William Greiman (the author of the SdFat library for Arduino) made me look a little deeper at what they were doing the rest of the time:

“Performance of microSD cards is not easy to predict on Arduino.  Arduino uses SPI to access the cards and the SD library has a single 512 byte cache. Modern SD cards are designed to be used on the high speed 4-bit SDIO bus with very large (16 KB or larger) multi-block writes and reads so they must emulate the single block access that Arduino libraries use.  This can mean much internal data movement, erasing of flash, and rewriting of data.”

This shows up clearly during data save events:

These screen captures of IDE serial potter output show current drawn during a data writing event with 256mB (left) and 2GB (right) Sandisk SD cards.  4kB of CSV format ASCII data was saved, and the gaps between the writing spikes were caused by i2c coms while SDfat’s cache was filled with data retrieved from the Eeprom on the RTC module. (click to enlarge)

After seeing that I tested a variety of different SD cards, finding that save event power use increased by more than 5x over the range from old 64 & 128mb Nokias to newer 2 & 4Gb Sandisk cards. 

It took me a good while to realize that I had fallen into yet another forest-for-the-trees situation, because even the worst offenders were only using ~30 mAs per save event, but all the cards were delivering similar sleep currents.  A day has 86,400 seconds in it, so the best sleepers, coming in around 70μA, were still burning six thousand milliamp seconds per day overall…

That brought me back to the question of de-powering those SD cards. I had been discouraged from trying this early in the project by some of Grieman’s other forum remarks where he suggested that there was nothing in the default SD library to support multiple shut downs & restarts safely.  But over time I found that Nick Gammon, and several others had card power control working with SdFat, and seeing the folks at OSBSS claim they had power cycled SD cards more than a hundred thousand times, was really the final straw.

As I had no logic level P-channel fets lying around I went with a garden variety 2n2222 BJT, configured as a ground side switch with a 30k pulldown. Driving it to saturation using a 330Ω base resistor (assuming Hfe = 30) should give me enough wiggle room to handle 150mA spikes, though it will burn 10mA to keep the BJT on for full second I needed to wait before pulling the plug.  Write latencies for SD cards can be quite large, and some cards have more than one stage of sleep, drawing around 1.0 ma for maybe a second before entering deep sleep.  But with 6000 mAs/day on the other side of the scale, I could afford the extravagance.

The cross leakage stuff I’d seen on EEVblog convinced me that I needed to actively pull up all of the SPI pins after the ground was disconnected. I cobbled together a set of  ON/OFF functions with pinmode commands, but it did not work reliably until I switched over to port manipulation (like they did at OSBSS), so the lines were all pulled simultaneously.  I was already disabling peripherals like the ADC with the PRR register, but that was just to save a little runtime power. Now it was required because when SPI is active, it controls MISO,MOSI & SCLK.  So you must shutdown the SPI interface before you can set those pins directly.

#include <LowPower.h>
#include <avr/power.h>
#include <SPI.h>
#include <SdFat.h>
SdFat sd;                                   // Create the objects to talk to the SD card
SdFile file;
const byte slaveSelect = 10;   // sd card slave select on pin D10
#define SDpowerPin 9            // pin controlling the BJT on the ground line for the SD card
boolean SDcardOn = true;     // flag for error routines
byte  keep_SPCR;
// spacer comment for blog layout
void setup () {
keep_SPCR=SPCR;                  // save the default SPCR register contents
. . . }

void turnOnSDcard() 

 {
pinMode(SDpowerPin, OUTPUT); digitalWrite(SDpowerPin, HIGH); //turn on the BJT on SD ground line
delay(6);                                            // let the card settle
// some cards will fail on power-up unless SS is pulled up  ( &  D0/MISO as well? )
DDRB = DDRB | (1<<DDB5) | (1<<DDB3) | (1<<DDB2); // set SCLK(D13), MOSI(D11) & SS(D10) as OUTPUT
// Note: | is an OR operation so  the other pins stay as they were.                (MISO stays as INPUT) 
PORTB = PORTB & ~(1<<DDB5);  // disable pin 13 SCLK pull-up – leave pull-up in place on the other 3 lines
power_spi_enable();                      // enable the SPI clock 
SPCR=keep_SPCR;                          // enable SPI peripheral
delay(10);  SDcardOn = true;       // just a flag
}

void turnOffSDcard() 

 {
delay(6);
SPCR = 0;                                         // disable SPI
power_spi_disable();                     // disable SPI clock
DDRB &= ~((1<<DDB5) | (1<<DDB4) | (1<<DDB3) | (1<<DDB2));   // set All SPI pins to INPUT
PORTB |= ((1<<DDB5) | (1<<DDB4) | (1<<DDB3) | (1<<DDB2));     // set ALL SPI pins HIGH (~30k pullup)
// Note: you must disconnect the LED on pin 13 or you’ll bleed current through the limit resistor
LowPower.powerDown(SLEEP_1S, ADC_OFF, BOD_OFF); // wait 1 second before pulling the plug!
delay(6);
pinMode(SDpowerPin, OUTPUT); digitalWrite(SDpowerPin, LOW);  //turn off BJT
delay(6); SDcardOn = false;


These two functions book-end any code that needs to write data to the SD cards:

turnOnSDcard();    flushEEpromBuffer();    turnOffSDcard();

All data saving functions start by checking the main battery to make sure there is enough power to save the data without a brown-out, and then re-initialize the cards with sd.begin before opening any files:

vBat = readBattery();   // This function shuts down the logger if the main battery is below 3.65V
if (!sd.begin(chipSelect, SPI_FULL_SPEED)) {
Serial.println(F(“Could NOT initialize SD Card”));Serial.flush();
error();  // note: the flag triggers:  if (SDcardOn) {turnOffSDcard();} in error function
}
delay(10);
file.open(FileName, O_WRITE | O_APPEND);  //see this post by Grieman
//…save your stuff…
file.close();


Looking at the datasheets for the Mic5205, or the Mcp1700,  you see the regulator dropouts can reach 300mV at 100mA+ currents you could see during SD card initialization, so your input cutoff for a 3.3V system needs to be above 3.65V to handle the load.   After the data is saved it is critical that all open files are closed properly before the turnOffSDcard function gets called, otherwise your data will be lost. The graphs tell me that a full one second delay before powering down the card is probably longer than it needs to be,  but in data logger applications it’s pays to err on the side of caution. According to Greiman:

“The standard says reliably removing power is not supported in SPI mode. It does suggest that you can remove power one second after the card goes not busy but does not guarantee this will work. You can’t depend on isBusy() to power down a card. It only means the card can accept a command. It may still be programming flash or moving data for wear-leveling. You really need the one second delay after not busy.”

Lately I’ve been using these 60¢ SD adapters, and removing the bottom three 10k smds that these boards have on the SCLK, MOSI & MISO lines. (the other resistor keeps the ‘RSV’ pins from floating) Having a pullup on the clock line wasted power during mode o sleep as the clock idles low, but now that I’m cutting power rather than just sleeping the SD cards, I could leave those resistors in place…then I wouldn’t need to pull up those lines in the code, (though I’d still have to pull SS…)

Of course, it was pretty flakey the first few times I tried it. Half of the loggers worked, but the other half were restarting every time there was a data save event (killing off SD cards in the process…) This problem affected every logger built around the Rocket Scream Ultra, which has been one of my favorite small form factor boards.  Closer examination of the two-penny clones that were working ok revealed that they had 10μF tantalum capacitors beside the voltage regulator rather than the little 1μFs beside the Ultra’s MCP1700.  So those cards were hitting the rails pretty hard when the SD ground line was re-connected, and this caused brief transients that were low enough to restart the processor on half of my units. Some add a small (33Ω) resistor in series to limit these inrush currents, but I found that adding 2-3 10μF (106) ceramics to buffer that spike got them all working ok, and for field deployment units I’ll probably add more.

I set a several units running on the bookshelf, with a rapid six second sampling interval. A couple of weeks later they were all still going, with some of them seeing more than 30,000 SD card power cycles without error. Given that the loggers normally see less than one save event per day, I’m calling that a successful test. If you run into issues, the first thing to try is extending that delay after sd.begin() and adding a few more delays throughout your functions. If you look at the spec you find that SD cards are allowed to take huge amounts of time for everything from initialization, to file open/close. While I did not see that in cards I used for my tests, these latencies are ‘officially’ allowed to stretch well beyond 100ms.

With both pin-powering on the RTC, and ground line switching on the SD card, the loggers get down to between 0.03-4mA between samples, which should push my operating lifespan into multi-year territory. Or, if I’m really lucky, they’ll make it through one winter-time deployment in Canada 🙂

I was also pleased to discover that the On/Off code seems to work on loggers that do not have the ground side switch installed provided I do not try to re-initialize the cards with sd.begin.  SPI shutdown & line pullup seems to cause the SD cards to enter sleep mode more quickly than they did before, and I have not seen any current leakage. So hopefully I won’t have to maintain vastly different code versions for older non-switched loggers. (Update 2017-06-12: Further tests of SPI shutdown, without the BJT to disconnect power from the SD card have not been reliable. Some worked, some didn’t. When I figure out why that is I will post an update)

Addendum 2017-06-06

A commenter over at Dangerous Prototypes made a point about my use of the 2n2222 which is important enough that I should pass it on:

“I’m surprised he didn’t check the 2N2222. Look at its data sheet, the V(CE) performance is not great. Take 0.3V at 100mA, then the SD card would have been actually running at 3.0V, right at the -10% VCC rating edge. I’m surprised the problems are not worse. Of course it would be extremely sensitive to VCC sag…”

The drop across the collector-emitter was something I had simply missed, and I still struggle to read those datasheet graphs properly.  And I was so used to seeing card operating voltage specified between 2.7-3.6v, that I also missed the fact that in SPI mode, only 3.3v is officially supported. The net result is that I’m probably sailing closer to the wind here than I realized, and I’m going to call this technique “experimental” until I see real-world deployments saving more than a year of data safely. And if I stay with ground-side switching in future, I will start looking for a good logic level N-channel  MOSFET, with low on resistance, to replace that BJT. The Supertex TN0702 looks like a good option with the promini’s with 3.3v logic.

Addendum 2017-06-06

Just thought I should post a reminder to test your SD cards thoroughly before embarking on SD power shut down experiments. I use SD formatter v4.0 & H2testw.  H2t is also a good way to make sure that you are not damaging your cards over time…

Calibrating Thermistors on a 3.3v Arduino

Selecting a thermistor (& series resistor) value

Most of the material you find on thermistors makes the assumption that you are trying to maximize sensitivity and interchangeability. But oversampling gives you access to enough resolution that sensitivity is less critical, and interchangeability only makes sense if you are putting them in a product with good voltage regulation. In that case, precision thermistors like the ones from US sensor are a good option, but according to Campbell Scientific, that choice has other knock-on implications:

“The resistors must be either bought or selected to 0.02% tolerance and must also have a low temperature coefficient, i.e. 10 ppm or preferably 5 ppm/°C.”

Like many better quality components, these resistors are often only available in SMD format, with minimum order quantities in the thousands. If you use a typical 1% resistor with a T.C. of 50 ppm or more, you could introduce errors of ±0.1°C over a 50°C range, which defeats the point of buying good thermistors in the first place.

Still, if I was only building a few sensors, I’d spring for the good ones. But now that I have oversampling working on the Arduino, I’d like to add a thermistor to every logger in the field, and the mix of different boards already in service means I’ll have to calibrate each sensor/board combination. That time investment is the same whether I choose a 10¢ thermistor or $10 one.

Power consumption is also important, making 100kΩ sensors attractive although I couldn’t even find a vendor selling interchangeable thermistors above 50k.  A low temperature limit of 0°C (the units are underwater…) and putting 1.1v on aref to boost sensitivity,  requires a 688k series resistor, which is far from the 1-3x nominal usually recommended:

Here I’ve overlaid an image from Jason Sachs excellent thermistor article at Embedded Related, which shows I will only see about ⅓ of the sensitivity I would get if I was using a 100k series resistor. I highly recommend reading Jason’s post, despite the fact that I’m ignoring almost all of his good advice here…  🙂

Using the internal band-gap voltage as aref improves the ADC’s hardware resolution from 3.22mV/bit to 1.07mV/bit.  This trick gives you a extra bit of precision when you use it at the default 10bit resolution, and I figured I could do it again to compensate for the lost sensitivity due to that big series resistor.

In return, I get a combined resistance of at least 700k, which pulls only 4.7μA on a 3.3v system.  Such low current means I could ignore voltage drops inside the processor and power the divider with one of Arduino’s digital pins.  In practical terms, burning less than a milliamp-second per day means adding a thermistor won’t hurt the power budget if I leave it connected to the rails all the time; which you can only do when self-heating isn’t a factor.  This is quite handy for the bunch of old loggers already in service out there, that I want to retrofit with decent temperature sensors. 

Even 100 ohms of internal chip resistance would produce only 0.5mV drop,  so depending on your accuracy spec,  you could use 16-channel muxes to read up to 48 thermistors without worrying about cable length.  There aren’t many of us trying to connect that many temperature sensors to one Arduino, but using a 100k  thermistor also makes me wonder if you could mux a bank of different series resistor values, pegging the divider output at it’s maximum sensitivity over a very large temperature range.

What is a reasonable accuracy target?

Combining 5¢ thermistors & 1¢ metfilms, means my pre-calibration accuracy will be worse than ±1°C.  Cheap thermistor vendors only provide nominal & βeta numbers, instead of resistance tables, or a proper set of Steinhart-Hart coefficients. So I might be limited to ±0.4°C based on that factor alone.  And it took me a while to discover this, but βeta values are only valid for a specific temperature range, which most vendors don’t bother to provide either.  Even with quality thermistors, testing over a different temperature range would give you different βeta values.

In that context, I’d be happy to approach ±0.1°C without using an expensive reference thermometer.  Unfortunately, temperature sensors in the hobby market rarely make it to ±0.25°C.  One notable exception is the Silicon Labs Si7051, which delivers 14-bit resolution of 0.01°C at ±0.1°C.   So I bought five, put them through a series of tests,  and was pleasantly surprised to see the group hold within ±0.05°C of each other: 

Temps in °CCompared to what I usually see when I batch test temperature sensors, this is pretty impressive for an I2C chip that only cost $9 on Tindie.

Ideally you want your reference to be an order of magnitude better than your calibration target, but given the other issues baked into my parts, that’d be bringing a gun to a knife-fight. 

So my calculations, with oversampling, and the internal 1.1v as aref become:

1) MaxADCReading                  (w scaling factor to compensate for the two voltages)

= ( [2^(OverSampledADCbitDepth)] * (rail voltage/internal aref) ) -1

2) Thermistor Resistance        (w series resistor on high side & thermistor to GND)

= Series Resistor Value / [(MaxADCReading / OverSampledADCreading)-1]

3) Temp(°C)                                  (ie: the βeta equation laid out in Excel)

=1/([ln(ThermResistance/Tnominal R)/βeta]+ [1.0 / (NomTemp + 273.15)]) -273.15

Seeing the error in my ways

I knew that the dithering noise would have some effect on the readings, and all the other source ADC of error still apply.  Switching to 1.1v reduces the absolute size of most ADC errors, since they are proportional to the full scale voltage. But the internal reference is spec’d at ±0.1v; changing the initial (rail voltage/aref voltage) scale factor by almost 10%.  Since all I needed was the ratio, rather than the actual voltages, I thought I could address this chip-to-chip variability with the code from Retrolefty & Coding Badly at the Arduino.cc forum.  This lets Arduinos read the internal reference voltage using the rail voltage as aref.

I started testing units in the refrigerator to provide a decent range for the calibration:

Si7051 in blue vs 100K thermistor in red. The sensors were held in physical contact. ADC was read with 1024 oversamples providing approximately 15bit resolution. Temps in °C.

and strange artifacts started appearing in the log.  The voltage readings from both the main battery and the RTC backup battery were rising when the units went into the refrigerator, and this didn’t seem to make sense given the effect of temperature on battery chemistry:

Si7051 temp. in °C on the left, with the RTC backup battery (V) in green on the right axis. The CR2023 is monitored through a 2x10MΩ divider, using the 3.3v rail as aref. The large number of ADC readings needed for oversampling has the side benefit that it lets you read very high impedance dividers, but by the time you reach 10Meg ohms, you pick up 5-10 points of noise in the readings. Which is why that coincell voltage line is so thick.

I think what was actually happening was that the output from the regulator on the main board, which provided the  ADC’s reference voltage for the battery readings, was falling  with the temperature.

When I dug into what caused that problem, I discovered that temperature affects bandgap voltages in the opposite direction by as much as 2 mV/°C.  So heating from 0°C to 40°C (and some loggers will see more than that…) reduces the 328P’s internal reference voltage by as much as a tenth of a volt. In fact, bandgap changes like this can be used to measure temperature without other hardware.  This leaves me with a problem so fundamental that even if I calculate S&H constants from a properly constructed resistance table, I’d still be left with substantial accuracy errors over my expected range.  Argh!

Becoming Well Adjusted

These wandering voltages meant I was going to have to use the internal voltmeter trick every time I wanted to read the thermistor.  It was mildly annoying to think about the extra power that would burn, and majorly annoying to realize that I’d be putting ugly 10bit stair-steps all over my nice smooth 15bit data. This made me look at that final temperature calculation again:

Temp(°C) =
1/([ln(ThermResistance/Tnominal R)/βeta]+ [1.0 / (NomTemp + 273.15)]) -273.15

which I interpret as:

 =fixed math(  [(ADC outputs / Therm. nominialR ) / Therm. βeta]  + (a #) ) – (a #)

Perhaps tweaking the thermistor’s nominal value (which I only know to ±5% anyway) and changing the (fictional) βeta values would compensate for a multitude of sins; including those voltage reference errors?  Then I could just pretend that (rail/aref) scaling factor had a fixed value, and be done with it:         (click image to expand)

So in my early tests, all I had to do was adjust those two constants until the thermistor readings fell right on top of the reference line.  Easy-peasy!

Well …almost. Repeat runs at 15bit (1024 samples) and 14bit (256 samples) didn’t quite yield the same numbers.  Applying the best fit Nominal and βeta values obtained from a 15bit run to 14bit data moved the thermistor line down by 0.05°C across the entire range (and vice versa). So the pin toggling method I used to generate the dither noise introduces a consistent offset in the raw ADC readings.  While that doesn’t completely knock me out of my target accuracy, I should generate new calibration for each oversampled bit depth I intend to use. It’s still good to know that the dithering offset error is consistent.

Throwing a Big Hairy Fit

I was pleased with myself for the simplicity of the Nominal/βeta approach for about two days; then I pushed the calibration range over 40° with a hot water bath:

Blue=Si7051 , Orange = 100k NTC thermistor.  1024 oversamples = ~15bit. Temps in °C.

This gave me targets at around 40, 20 and 5°C.  But no combination of Nominal & βeta would bring all three into my accuracy range at the same time.  Fitting to the 20 & 40 degree data pushed the error at 5°C beyond 0.2° :             (click image to enlarge)

…and fitting to 20 & 5, pushed the 40C readings out of whack.  After more tests I concluded that tweaking βeta equation factors won’t get you much more than 20° of tightly calibrated range. 

My beautiful plan was going pear-shaped, and as I started grasping for straws I remembered a comment at the end of that Embedded Related article

“… in most cases the relationship between voltage divider ratio and temperature is not that nonlinear. Depending on the temperature range you care about, you may be able to get away with a 3rd-order polynomial or even a quadratic..”

Perhaps it was time to throw βeta under the bus, and just black-box the whole system?   

To find out, I needed to prune away the negative temperature regions where the voltage divider had flat-lined, and remove the rapid transitions since the thermistor responds to changes more quickly than the si7051:                 (click image to inflate)

Then it was time for the dreaded Excel trend line:

Ok, ok. I can hear people inhaling through their teeth from here. But with 15 sigfigs, Excel seems like the height of luxury compared to the constraints in μC land.  I wonder what an advanced modeler like Eureqa would have produced with that dataset? 

The trick for getting workable constants is to right-click the default equation that Excel gives you, re-format it to display scientific notation, and then increase the number of displayed digits to at least six.  

Some people use the LINEST function to derive these polynomial constants but I’d advise against it because seeing the raw plot gives you a chance to spot problems before you fit the curve. When I generated the first Temp vs ADC graph, the horizontal spread of the data points showed me where the thermistor and the reference thermometer were out of sync, so I removed that data.  If I had generated the constants with =LINEST(Known Y values, X values^{1,2,3,4})  I could have missed that important step.

For the following graphs, I adjusted the trend line to display to nine insignificant digits:     

Blue =Si7051 reference, Orange is that 20&40 best fit from tweaking Nominal & Beta values, and the yellow line is the 4th order polynomial from Excel.   Temps in °C. (Click to embiggen)

It took a 4th order polynomial to bring the whole set within ±0.1° of the reference line and 5th order did not improve that by much.  Now I really have no idea where the bodies are buried!  And unlike the βeta equation, which just squeaks in under the calculation limits of an Arduino, it’s beyond my programming ability to implement these poly calcs on a 328 with high bit depth numbers. I certainly won’t be writing those lunkers on the bottom of each logger with a sharpie, like I could with a pair of nominal/βeta constants.

This empirical fit approach would to work for any type of sensor I read with ADC oversampling, and it’s so easy to do that I’ll use it as a fall back method whenever I’m calibrating new prototypes. In this case though, a little voice in my head keeps warning me that wrapping polynomial duct tape around my problems, instead of simply using the rail voltage for both aref & the divider, crosses some kind of line in the sand. Tipping points can only be predicted when your math is based on fundamental principles, and black-boxes like this tend to fail dramatically when they hit one.  But darn it, I wanted those extra 1.1v aref bits! Perhaps for something as simple as a thermistor, I’ll be able to convince the scientist in the family to look the other way.

Addendum 2017-04-28

Seeing that trend-line produce such a good fit to the temperature data, made me think some more about how I was trying to stuff those system side errors into the βeta equation, which just doesn’t have enough terms to cope.  By comparison, the Steinheart-Heart equation is a polonomial already, so perhaps if I could derive some synthetic S&H constants (since my cheap thermistors didn’t come with any…), it would peg that ADC output to the reference line just as well as Excel did?

I rolled the voltage offsets into the thermistor resistance calculation by setting the (rail voltage/internal aref) scale factor to a fixed value of 3, when in reality it varies from slightly below to slightly above that depending on the board I’m using:

1) MaxADCReading                  (w scaling factor to compensate for the two voltages)

=(2^(OverSampledADCbitDepth) * (3)) –1

2) Thermistor Resistance        (w series resistor on high side & thermistor to GND)

= Series Resistor Value / ((MaxADCReading / OverSampledADCreading)-1)

and I went back to that trimmed 40-20-5 calibration data to re-calculate the resistance values. Then to derive the constants, I put three Si7051 temp. & thermistor resistance pairs into the online calculator at SRS:

(Note: There are premade spreadsheets that you can download which will generate S&H constants, or you can build your own in Excel. There’s also coefficient calculators out there in C, Java, etc. if that’s your thing.)

With those Steinhart-Hart model coefficients in hand, the final calculation becomes:

3) Temp °C =1/( A + (B * LN(ThermR)) + (C * (LN(ThermR))^3)) – 273.15

and when I graphed the S&H (in purple) output against the si7051 (blue) and the 4th order poly (yellow), I was looking at these beauties:

and that fits better than the generic poly;  nearly falling within the noise on those reference readings. With the constants being created from so little data, it’s worth trying a few temp/resistance combinations for the best fit. And this calibration is only valid for that one specific board/sensor/oversampling combination;  but since I’ll be soldering the thermistors permanently into place, that’s ok.  I’m sure if I hunt around, I’ll find a code example that manages to do the S&H calculations safely with long integers on the Arduino. 

So even with cheap parts, oversampling offsets & bandgap reference silliness, I still made it below ±0.2°C over the anticipated temperature range.  Now, where did I put that marker…

Addendum 2017-04-27

Just a quick note to mention that you need to tape the thermistor to the si7051 sensor so they are held in physical contact with one another. The thermistors are tiny & react to temperature changes much faster than the si7051’s which have a much larger thermal mass because of the breakout board they are mounted on. So the temp/resistance pairs don’t match up as well as they could if the sensors are in physical contact with one another.

Addendum 2017-06-05

With 1.1v aref in the mix,  my 15bit oversampled resolution on those 100k thermistors varies between 0.002 and 0.004°  from 20-40°C. But I was throwing the bandgap aref in just to see if I could still make it work. From a calibration point of view, it’s better to better to use the rail voltage on aref, and remove that 3x ratio from the MaxADCReading calculation.  This will lower the resolution to somewhere between 0.006 to 0.012C with a 688k series resistor unless you bump up the oversampling to compensate. In addition to tripling my noise/toggle-pin current, how much extra power do I have to pay to get that resolution back if I’m using the 3.3v rail as aref?

In my oversampling experiments, I found that the Arduino ADC works well at 250 kHz, delivering just under 19230 ADC readings /second. For the purpose of estimation, assume the Pro-mini style boards I’m using draw about 5mA during the sampling time, and I take a reading every 15 minutes (= 96 readings per day) :

15bit= 1024 reads/19230 r/sec =0.053s*5mA =0.26 mAs*96/day=~ 25 mAs/day
16bit= 4096 reads/19230 r/sec = 0.213s*5mA =1.00 mAs*96/day= ~102 mAs/day
17bit= 16384 reads/19230 r/sec = 0.852s*5mA =4.26 mAs*96/day= ~408 mAs/day

so it would cost me another 385 mAs/day to reach a resolution slightly better than I was achieving with the 1.1v bandgap on aref. Given that a typical AA battery runs about 2000 mAh = 2000 mAh*3600 sec/hour =~7,000,000 mAs, it would be quite a while before that breaks the power budget.  Removing the ratio dependency also means that your S&H constants are for the resistor/thermistor pair only, making that calibration independent of what system you connect them to.

Using an Rnominial=100k series resistor would give about the same effective resolution boost as going to 17 bit, but that option costs you more power if you are leaving the thermistor powered all the time:

3.3v / 780k combined resistance  = 4.23μA x 86400 sec/day  = 366 mAs/day
3.3v / 200k combined resistance  = 16.5μA x 86400 sec/day  =  1425 mAs/day

You can power the thermistor from a digital pin, but since I’m already using digital-pin toggling to generate noise for the oversampling, I still need to test if I can combine pin power for the sensor with my oversampling technique. It’s possible that the thermistor bridge needs to be powered by the more stable rails, while I’m shaking aref inside the processor, because if the voltage on the divider started moving in sync with the ADC noise, the dithering noise will effectively disappear, and my oversampling would stop working.

Even before doing this test, I have a sneaking suspicion that 100k series vs. oversampling vs. other techniques  will end up converging on the same effective resolution in the end. And I’ll even hazard a guess that the point of diminishing returns is somewhere around 0.001°C, since that’s what you see offered by quite a few high-end temperature loggers.

Addendum 2017-09-24

Just posting an update about pin-powering the thermsitor dividers while using the 3.3v rail as aref: everything works, but as I suspected you need to stabilize the thermistor with a small 0.1uF capacitor or the dither noise vanishes.  This also requires you to take the RC time constant into account, waiting at least 5x T  for that parallel cap to charge before you start reading the divider. You can sleep the processor during this time, since I/O pin states are preserved.

Degree Celsius vs. Time with lines offset for easier visual comparison:  The blue line is over-sampled output from a pro-mini clone reading a 100k Thermistor /100k series voltage divider. Aref was set to the 3.3v rail, with a 100nF capacitor in parallel with the thermistor on the low side.  This RC combination has a time constant of ~10 milliseconds.  A 0.12 mA pin-current provided sufficient noise for dithering 1024 readings: to deliver an effective resolution of ~0.0028° at 24°C.  For comparison, the red line is the output from an I2C si7051 sensor on the same logger, with a resolution of 0.01°C.

So using a 100k series resistor with 3.3v aref really does deliver the same effective resolution as the 680k series/1.1v aref combination, and it does not suffer the problem of bumping into the aref voltage at a certain temp.  I’m using 100k termistors so the pin resistance (~40 ohms) will introduce less than 0.05% error over the range; though this pin-drop error would be higher for therms with lower Rnominal values.

Since I’m using cheap eBay 100k’s and a host of other no-name components, I have to calibrate each logger/thermistor/O.S. bit-depth combination.  This isn’t much of a burden for the overall workflow, since I always give new loggers a shake-down run, in fact, I usually do a fast sampling burn for at least a week before considering a unit ready for deployment:

That Degree vs Time image above was an excerpt from a calibration run like this. I’ve found that Freezer (morning)->Fridge (afternoon)->Room (overnight) is easier to manage than the reverse order, and gives enough time at each temperature to deal with thermal lag differences between the thermistors and the reference sensors.

As before, when I do the thermistor resistance calculation I make the assumption that everything in the system is behaving perfectly (which is obviously not true). So errors from things like pin drops, temp. coefficients, ADC gain, etc., are getting rolled into the S&H constants.  Essentially, I’m eliminating a host of irritating corrections in exchange for the interchangeability between sensors that I might have if I took all those factors into account. This makes it easier to standardize the code between loggers, and is a reasonable trade-off for loggers that I won’t be seeing again for several years, but if I have to swap some components at that time, I’ll need to do  another calibration.

Addendum 2017-11-05

Looks like Sensirion’s new STS35 has ± 0.1°C accuracy like the si7051 I’m currently using as a calibration reference.  Hopefully that shows up on Tindie soon.

Improving Arduino ADC resolution with Oversampling & Noise

Thermistors are really twitchy, so you need to put them inside a big lump of thermal inertia before you start.

The slightest breeze makes glass bead thermistors jitterbug like crazy, so put them inside something with a decent amount of thermal inertia before you do any oversampling experiments.

While I was figuring out how to read thermistors, I came across claims that you can improve the resolution of any Analog-to-Digital converter with a technique called oversampling & decimation. I had already doubled the number of ADC bits covering my target temperature range by powering a thermistor divider from the rails and using the internal 1.1v as the analog reference.  My gut feeling was that aref based ADC bits were somehow better than any I could synthesize, but I was still curious to see if I could add oversampled bits to the ones obtained by the bandgap aref trick.

At first bounce, the method appeared to be incredibly simple, to get n extra bits of resolution, you need to read the ADC four to the power of n times.  Generally you have to add three extra bits (43= 128 samples) to see approximately an order of magnitude improvement in your real world resolution. With thermistor dividers, you typically get about 0.1°C from the default ADC readings, and 128 samples bumps that to 0.012°C.  Taking (46= 4096) samples would bump that up to ~0.0015°C which, as the saying goes, is good enough for government work… 

I usually over-sample one power more than needed for my target resolution, so I’d use for four extra bits to be sure of that order of magnitude improvement, which requires the sum of 44= 256 readings:

uint32_t extraBits=0;    // use an unsigned integer or the bit shifting goes wonky
for (int k = 0; k< 256; k++) {
extraBits = extraBits +analogRead(AnalogInputPin);
}

which is then decimated by bit shifting right by n positions:

Oversampled ADC reading = (extraBits >> 4);

This combination lets you infer the sub-LSB information provided there is enough random noise in the signal for the lowest ADC bits to toggle up and down while you gather those readings. But without the noise, all of the original ADC readings are the same, and the oversampling technique does not work. It’s a good idea to keep in mind what you are aiming for with your oversampling. Generally you have to take enough samples for three extra bits (43= 128) to see approximately an order of magnitude improvement in your the real world resolution. With thermistor dividers, you typically get about 0.1°C from the default ADC readings, and (43= 128) samples gets you to 0.012°C.  Taking (46= 4096) samples would bump that up to 0.0015°C which, as the saying goes, is good enough for government work… 

To show you what that kind of failure looks like, here is oversampling & decimation being done over 4096 readings with no noise or dither signal applied to a 10k NTC thermistor divider read with 1.1v aref:

This is an example of oversampling with no dither signal being applied. So this is the nul result

These are readings from a 10k NTC thermistor divider, and I’ve offset these records from each other by 0.1° for easier comparison. The one-shot ADC readings of the thermistor bridge in purple are converted to °C, as are 4096 sample readings at the default 125kHz(ps64) in grey,  250kHz(ps32) in orange and 500kHz (ps16) in green. With such a large number of samples, the averaging produces some smoothing whenever the raw ADC readings near a transition point, but if you see “rounded stair steps” like this then oversampling is not working properly  the curves shown above are all FAILURES.

Some microprocessors have enough jitter in their readings to use oversampling technique with the natural toggling of the least significant bit.  A few brave souls have even tried to improve the AVR’s crude internal temperature sensor with the technique.  But most of the time, there is not enough naturally occurring noise, and you need to add a synthetic dithering signal to force those LSB’s to toggle.  This is mentioned from time to time in the forums, with a number of references to AVR121: Enhancing ADC resolution by oversampling, but I found frustratingly few implementations using an Arduino that were described in enough detail for me to replicate them.  Most of the technical docs were focused on audio applications, and I was quickly buried under thick mathematical treatments warning me not to interpret the Effective Number of Bits (ENOB) as Effective Resolution (what?), and describing a host of other quid pro quos like signal synchrony.

This is qwerty's original dither circuit from the freetronics forum post at: http://forum.freetronics.com/viewtopic.php?t=5589#p11126

This is Qwerty’s original dither circuit from the freetronics forum. If you are using an UNO, this works well. Of course the ratio between the 5v rails, and the internal bandgap reference,  means you also have extra ADC resolution available without oversampling if you use the 1.1v aref trick, but oversampling gives you more bits for your effort.

About the only useful thing I got out of most of those refs was the apparent consensus that any synthetic dithering signal needs to be at least 2x the voltage per bit on your ADC (although you can use a larger dither signal without causing problems) and triangular dither signals work better than natural noise.  But few of those references said anything about extending ADC resolution, as they were primarily focused on improving the ADC’s signal to noise ratio.

And then there was the fact that several of the older hands seem to dismiss the whole idea as not worth the bother because you had to add so much additional circuitry that using an external ADC was a simpler, cheaper approach.  In fact the subject triggered the closest thing to a flame war I’ve ever seen at the usually staid Arduino playground.  So I was about ready to give up on the idea when I came across a post by user QWERTY at the Freetronics forum explaining how he used a simple RC filter to turn an Arduino’s 480 Hz PWM output into a 9mv p-p triangular dither, which he patched directly into the center of a thermistor bridge.

Yes it is possible to add a jumper on the Aref line of a pro mini.

You can patch into the aref line on a Promini by soldering a jumper to the end of the little stabilizing capacitor.

Holy cow! A solution that only needed a few cheap parts and couple of pins. What the heck were those other guys gassing on about?   My first thought was to try to take the output from Qwerty’s RC filter, and put it onto the Aref as they did in AVR 121.  A compelling idea since putting the dither directly on aref means you don’t have to interfere with the sensor(s), and the same dither circuit would work for all of the analog inputs.  In addition, I was using large resistance voltage dividers to monitor Vbat without wasting power and the high impedance forced me to add a capacitor to feed the ADC’s sample and hold input.  I knew that low esr cap would kill any dither signal that was applied directly to the main battery divider.

fig35avr121

This L-P filter from AVR121 works too, but modifying the circuit to give you other aref voltages is a bit of a pain.

I tried many different combinations, but I never saw the voltage on aref that I expected.  It took ages to discover that ~32k of internal resistance gets connected when you place an external voltage on the aref line, and that forms a ‘hidden’ voltage divider with your circuit. Grrr…

I did eventually get a few of those circuits working, but that internal resistor  seemed  to be slightly different on each board I tried, and I didn’t know if it was going to be stable with temperature, time, etc.  Another important issue was that I was switching from the internal 1.1v aref to read the thermistor, back to using the default 3.3v for other readings during the logger operation. So to put the dither directly into aref meant I would also need some way to modify the baseline aref voltage on the fly.  

Tune the resistor ratio, and roll PWM2 duty cycle and I'm pretty sure this circuit form Open Labs would give you variable Aref voltages.

Tweak the resistors & this circuit could give you variable arefs AND dithering.

I suppose that a truly elegant solution would do that with a PWM/RC filter circuit generating a variable DC voltage, and using a second PWM input to add the much smaller dither signal.  You could tune the dithers pk-pk amplitude to match the adjusted LSB, by the way you varied PWM2’s duty cycle (or by using the tone function)  during the readings.  But I knew working that out would probably give me a host of other problems to resolve (esp. with timing) and I was after a simple solution, with the smallest number of parts.  So I eventually abandoned the “dither on aref” approach.

This brought me back to Qwerty’s method of putting the triangular dither signal on the center of the thermistor bridge. My first task was to change that RC filter to lower the 9mv swing on that 5v circuit to match the much lower 1.1mv/LSB you get when using the internal bandgap as aref.

The power supply ripple calculator at OKAWA Electric was a perfect tool for this job:

oklowpassfilter

3.6mV was just an arbitrary 'close enough' point for me to start at as I had those components in the parts bit already.  But if you see random flat spots in your oversampled readings at the default ADC speed, then try increasing the ΔV pk-pk of your dither signal a little bit.

3.6mV was just an arbitrary ‘close enough’ point for me to start at as I had those components in the parts bit already.  But if you see random flat spots in your oversampled readings at the default ADC speed, then try increasing the ΔV pk-pk of your dither signal a little bit.

…which revealed that a 4.7MΩ/0.1uF RC combination would take the 3.3v 480Hz PWM on D6 and bring it down to  ~3.6mv peak to peak.  I immediately  hopped over to the Falstad circuit simulator to see the see how this worked.  To simulate an Arduino’s positive PWM, I used a 3.3v square wave source with an offset of 3.3v.  The little 10nf coupling cap prevents the pins DC voltage from affecting the thermistor reading, and the 2k2 bridge resistor prevents the dither signal from being grounded out when the 10K NTC thermistor resistance gets very low.  One of the coolest features of this simulator is that if you build a circuit with it, you can export a web link (like the ones above) that rebuilds the circuit instantly, so you can compare different versions simply by keeping those links in your log.

rcrisetime_png

The RC settling time is shown on the Okawa calculator’s step response graph, or you can watch the voltage rise on the scopes in Falstad by restarting the simulation with the buttons on upper right.

I love using Falstad for “What happens if I do this?” experiments. Of course these usually fail, but in doing so they show me that there are things about a circuit that I still don’t understand.  One thing that gave me a lot of grief when I started working with these  dithering circuits was that I did not appreciate how much time they need to stabilize.  This gets worse if you start disconnecting the thermistor  divider to save power between readings.  

So although I was getting smoother curves, and resolution that looked about 10x better than my raw ADC readings:

excerpt from 1024 oversampled temp record on Arduino ADC with triangular dither , 100kthermistor

Here I’ve converted these 1024 sample curves to °C , and artificially offset each curve by 0.05° from the next to it for easier comparison. The one-shot 10bit ADC reading at the default 125kHz (ps 64) is in purple, with other ADC speeds:  250 kHz (ps32) in orange,   500 kHz (ps16) in green, and 1 MHz (ps8) in blue.

At the height of my coupling capacitor lunacy I produced this beast, thinking I could simultaneously read a reference bridge, and correct away any offsets.

At the height of my coupling capacitor infatuation I produced this beast, thinking that if I could simultaneously add dither to a reference bridge I would be able to correct away ADC offset & gain errors, along with the offset caused by the dither signal, at the oversampled bit depth. But all those capacitors added artifacts to the readings when I reconnected GND through that mosfet, producing weird spikes in the data if I took readings less than two minutes apart (?)

…in any set of successive readings, the offset between the oversampled readings and the one shot ADC reading was changing depending on how long the PWM had been running.  No problem I thought, I’ll just throw in another coupling cap to block that slowly rising DC voltage, and connect the ADC input on the thermistor side. Unfortunately replacing the 2k2 bridging resistor with a coupling capacitor forms a high pass filter with the thermistor itself, forcing you to increase the size of the cap to raise the filters cutoff frequency above your 480Hz PWM. But that increases your RC time constant so then the filter starts to act like a differentiator: distorting your nice triangular dither signal (see pg12 of this pdf), and in some cases even reverting it back to the original square wave you started with… Argh!

So the result of all that trial & error is the basic PWM->triangular dither method works well, but you have to wait for the RC filter’s output to stabilize or it messes with your accuracy. And you still end up with a small offset in the ADC readings of 1/2 your dither signals peak to peak, because the original PWM square wave can only be positive.

Crank it up

But no one wants to see a data logger burning away precious milliamp-seconds just twiddling its PWMs!  With guidance from Nick Gammon’s fantastic ADC page, I had already been messing around with pre-scalars to increase the temporal resolution of my UNO DAQ.  I was further encouraged by this line from AVR120    “For optimum performance, the ADC clock should not exceed 200 kHz. However, frequencies up to 1 MHz do not reduce the ADC resolution significantly.  …and there were some tantalizing hints that cranking up the speed might also increase the internal noise enough to make oversampling work better. 

To figure out how fast your ADC is running:

System clock / prescalar = ADC clock,  ADC clock /13 = # of ADC reads/second

The core clock speed on 3.3v promini style boards is 8 MHz, providing:

8 MHz / 64 = 125 kHz /13 ticks    = 9600 /sec      (256 reads =27.6ms, 1024 =106ms, 4096 =426ms)  (default) 
8 MHz / 32 = 250 kHz /13             = 19230 /sec     (256 reads = 13ms,  1024=53ms, 4096=200ms)
8 MHz / 16 = 500 kHz /13             = 38000 /sec     (256 reads = 6.7ms, 1024=27ms, 4096=108ms)
8 MHz /   8 = 1 MHz /13                 = 76900 /sec     (256 reads = 3.3ms, 1024=13ms, 4096=53ms)

Your sensors output must be stable while you gather these samples and this limits what kind of phenomenon you can measure. At the default ADC clock speed, trying to add six extra bits of resolution (46 = 4096 readings) means you can only capture about 2 samples per second. That’s pretty darned slow for data acquisition! In fact, it’s so pokey that some people implement ring-buffer schemes to provide access to an oversampled reading at any time, without having to grab a whole new set of samples. A neat trick if you are continuously monitoring a sensor that changes slowly, and you have enough memory to play with.  Given the powers of 4 relationship between the different bit depths, it’s easy to see how you might hop-scotch through shorter 64 sample readings, and then combine those into a sort of rolling average version of a 256 sample reading if you don’t have quite enough ram for the full ring buffer approach.

enobs

My tests agree with the results posted at Open Labs. You can only push the ADC clock so far before you lose hardware bits, and this defeats the resolution gained from oversampling by making your accuracy worse. You can see this effect in the 1MHz line in the previous 1024 sample graph. Most AVR’s are lucky to get above 9 ENOB’s at their default settings.

200 kHz is the ‘official’ ADC speed limit for 10 bit accuracy, but I didn’t see any  significant difference between oversampled readings taken at the default 125kHz clock (ps 64), and those taken at 250kHz (ps 32).  At 500kHz (ps 16) the readings were good most of the time, but during rapid temperature transitions the readings started to ‘wiggle’ as though the dither signal was occasionally dropping out.   At 1MHz (ps 8) the curves wander around quite a bit, and I was seeing errors of ±0.05°C or more with some prolonged flat spots starting to appear. What’s interesting about this is that the triangular dither RC filter puts a capacitor across the thermistor, which should reduce the input impedance seen by the ADC and allow for faster readings.  But this did not reduce the 500kHz wiggle / 1MHz wandering in any of my test runs.  The ATmega328P datasheet quotes 2 LSB’s (typical) of absolute accuracy with an ADC clock at 200 kHz, but 4.5 LSB’s (typical) at an ADC clock of 1 MHz, and there is no point in pushing clock speeds if the accuracy gets worse by that much in the process.

So you can always double the ADC clock speed for oversampling, but going up to 500kHz depends on whether you can live with the accuracy errors that prescalar creates.  Those 500kHz wiggles become less evident as you progress from 256, to 1024, to 4096 readings, but that’s probably just an artifact of the smoothing.  The other thing to keep in mind is that one full cycle of the 480Hz PWM takes  ~2 milliseconds, but 256 readings at a 500kHz ADC clock takes only 6.73 milliseconds – so there is a high probability that dither signal synchrony issues creep in at the higher ADC speeds to produce offsets that affect the entire curve. Ideally you’d want the time you spend gathering the over-samples to be an exact multiple of the dither cycle time…

Let’s make some noise!

Hotter prescalars cut the oversampling time down dramatically, but I could not see how to avoid that RC settling time, which seemed to require about 50-60ms of PWM operation before the offsets became tolerable.  So I went back to the proverbial drawing board and asked myself, what if forget about the triangle dither signal, and try oversampling with some sort of random noise?

The first hurdle there was:  How was I going to generate this noise if the processor was already busy taking ADC readings?  The beauty of PWM based dither is that it just chugs away in the background, leaving the processor free.  As usual, Nick Gammon provided an elegant solution to this problem with code on his page about interrupts which showed how to read the ADC asynchronously:

// Note: Before calling this function, I change to the internal 1.1v aref and set the ADC prescalars
// but you can leave them at the defaults: see: https://www.gammon.com.au/adc for more details
volatile int adcReading;
volatile boolean adcDone;
boolean adcStarted;
unsigned int  adc_read;

unsigned long asyncOversample(int readPin, int extraBits)

    {
int i=0;int j=0;
int var=256;                                  //default is 4bits worth of oversampling
if(extraBits == 5){var=1024;}
if(extraBits == 6){var=4096;} //I’ve only included three options here, but hopefully you see the pattern
unsigned long accumulatedReading = 0;
adc_read=analogRead(readPin);   // a throw away reading to connect the ADC channel
//delete me:  simply as spacer
pinMode(5, OUTPUT); digitalWrite(5, LOW);  // set the pin you are toggling to OUTPUT!
//delete me:  simply a spacer a spacer comment for blog layout
while(i < var){    // asynchronous ADC read from  http://www.gammon.com.au/interrupts
  if (adcDone)
  {adcStarted = false; accumulatedReading += adcReading; adcDone = false;i++;}
  if (!adcStarted)
  {adcStarted = true; ADCSRA |= bit (ADSC) | bit (ADIE);}

  PORTD ^= B00100000;  // XOR toggle D5 w green LED & 30k limit resistor (see  below for details)
}   // end of while (i < var)

pinMode(5, INPUT);digitalWrite(5, LOW);  //turn off the toggle pin
if(extraBits == 4){accumulatedReading=(accumulatedReading >> 4);}  // Decimation step for 4 extra bits
if(extraBits == 5){accumulatedReading=(accumulatedReading >> 5);}  // 5 bits
if(extraBits == 6){accumulatedReading=(accumulatedReading >> 6);}  // 6 bits
return accumulatedReading;
}   //end of asyncOversample function

ISR (ADC_vect)     // ADC complete ISR needed for asyncOversample function  
  {  adcReading = ADCL | (ADCH << 8);adcDone = true; }

 

Next I had to generate the noise itself. People use Zenner diode breakdown to produce random number seeds, but thought I would see if I could generate noise inside the processor, since there seemed to be no end of people complaining about the Arduino’s ADC in the forums. However when I actually tried to do this by connecting pull-ups,  changing I/O settings, an every other kind of processor toggle I could think of, I got nothing.  That ADC was solid as a rock until I started flipping the pins connected to the external indicator LED.   Even then, the early results were wildly inconsistent, with the same code producing good oversampling on one unit, but not another.

Like the hidden resistor problem, it took me a while to notice that the random bunch of LEDs on my breadboard test units had significantly different forward voltage drops from one LED the next, and from one RGB color channel to the next.  Once I realized how much that was affecting the results,  it didn’t take long to determine that that the noise generating sweet spot was somewhere around 0.04mA of pin current:

An example of oversampling with pulsed pin current of 0.038mA to generate ground line noise.

One-shot ADC reading shown in purple, with oversampled readings taken at 125kHz (ps64 default)  in grey, 250kHz (ps32) in orange, 500kHz (ps16) in green. All readings are converted to °C, and I’ve offset these curves for clarity, as they would otherwise be on top of one another. You can clearly see the PS16 wiggle as the temperature falls, and the sharp eyed will notice there are still offsets between the different runs which were all taken in quick succession. These seem to be more apparent in the longer slower oversampling runs than they are in the the shorter faster ones… darn it…

Unlike triangular dither techniques, which will tolerate a fairly large ΔV, this noise based method stopped working (ie: flat spots started appearing) when the toggled pin current went below 0.02mA, and the curves became pretty scratchy above 0.06mA  indicating there was too much noise.  That’s a fairly tight range, and it was sheer luck that the 30k limit resistor I was using on my indicator LED’s brought me close enough to spot the effect.  So my current target is ~0.04mA of pin current for 1.1v dithering. And there was nothing special about the LED being there either, as tests using a simple 82.5KΩ resistor from the  PORTy ^= _BV( PDx/PBx );    toggled pin to ground produced good results.  This is pure conjecture on my part, but if you assume the mosfets on the I/O pins have about 40Ω of internal resistance with 3.3v control, then 0.04mA pin current would produce a voltage drop of ~1.6 mv – which is suspiciously close to the 1.1mv/LSB resolution of the ADC with the internal bandgap set as aref.  That puts this dither noise right into the 1-2x voltage per bit recommendation from the literature.

rtcdividerreadings

Here I’m oversampling with 1024 readings from a 2x10MΩ divider which cuts the voltage of the RTC’s backup coin cell in half. 250kHz (psS32) in orange, and 125kHz(ps64) in grey. These are the raw readings with aref set to the default 3.3v and there is no capacitor on the divider. This is far beyond the 10k input impedance the ADC was designed for, but I think the many repeated readings you do with oversampling helps the 14pF sample&hold caps do their job. At this resolution, the CR2032 seems to be acting like another temperature sensor (?) UPDATE: So this actually was the battery responding to temperature rather than the dithering method, which does not work with the rail voltage on aref unless you add a cap to the voltage divider.

This pin-toggling noise technique is not exactly a one size fits all solution, and the exact current required to induce ADC bit toggling will vary depending on which board you are using, and especially on which capacitors are being used smooth the output from the voltage regulator.  So you will have to noodle around a bit to find the correct resistor value to use for your particular Arduino. I’d start with a resistor value that draws enough current to give you a voltage drop on the digital pin mosfet to approach 2x your ADC’s mV/LSB resolution. With 3.3v as aref (so 3.22mV/bit), I would use a pin resistor of  about 27.5k to generate a pin current of 0.12mA, and a pin vdrop of ~4.8mV.   With 5v control logic, the mosfets controlling the digital pins are more fully turned, so the pin resistance is somewhat lower; around 25-30 ohms. With 5v on aref your resolution is about 4.88mv/bit, and the dither resistor would have to pull around 0.39 mA to shake the rail with a vdrop twice that mv/bit, so the dithering resistor would need to be somewhere around 12.8 kΩ.

On new builds I will measure the forward voltage drop of the indicator LEDs and change the limit resistor to give me the current through those I need to generate dither noise. That way I don’t need to any new digital lines for the oversampling process, though this will entail checking every LED, as there is significant vf variation between batches.  The blue channel on the RGB’s I have lying around have a vf of ~2.473v, so 0.827v will be left for the resistor to cover with a 3.3v rail.  To achieve a target pin current of 0.12mA the limit resistor would have to be 0.827v/0.00012mA = 6.89kΩ.

This method is also critically dependent on the tiny capacitor stabilizing the aref voltage. When I tried it on the units I had left over from the ‘dither on aref’ experiments, the method did not work at all if the aref stabilizing capacitor had been removed.  I also suspect that the voltage on the capacitor ‘adjusts’ to the noise pulses over time, which might be causing the 0.02C difference between the 256 & 1024 readings shown above. So there could be another settling time issue if you take a large number of over-sampled readings in rapid succession. Larger caps stabilizing the rail voltage on breakout boards may also affect the method.

This technique will work with any resistive sensor being read with a simple voltage divider, provided there are no capacitors nearby to smooth out the noise which is vital for oversampling to work.  I’m not going to pretend to understand all the math behind it,  but it’s probably safe to say you can add somewhere between 2-5 extra bits of resolution to your ADC before the technique suffers from other limiting problems.  Although the 256 sample curves are a bit gritty, you can make that many samples with the ADC clock at  250kHz in ~13milliseconds, which doesn’t impact the power budget too much. If something interesting starts happening, you can dynamically enable another bit or two of resolution in the code to zero in on it.

Overall, I’d say the results from oversampling with toggled-pin noise are not quite as smooth as the curves you can get with a well tuned triangular dither, but I’m happy to trade that last bit of synthetic resolution for a method that’s instantly available for all of the ADC inputs with less chance of a synchrony problem.  The icing on the cake is that I don’t have to add any extra circuitry to use oversampling on the fleet of loggers already on deployment, because all I have to do is toggle the indicator LEDs they have on board since their limit resistors were already in the current range I need…YES!

Addendum 2017-04-26:

I’ve moved on to calibration, and in the process I learned that regulator & bandgap voltages change a fair bit with temperature. So it’s probably not a good idea to use the internal bandgap on aref with this oversampling method if you want thermistors calibrated over a wide temperature range. But I did it anyway.

In those tests I used a 688k series resistor with a 100k thermistor, so I was far from divider’s optimum of Rseries=RTnominal. I was taking 1024 oversamples, adding five oversampled bits to ADC, and I was using the internal bandgap voltage on aref, which added another bit.  Since I was on the tail end of the divider sensitivity curve, the effective resolution changed quite a bit over the range. The output shifted from ~0.0018°C/bit at 20°C, to about 0.0038 °C/bit up at 40°C, and I will have to do some homework to figure out how much of that signal is being lost to noise. This is better resolution than some people achieve reading thermistor bridges with the 16bit ADS1115, though gathering all those readings means I can only capture 18 samples per second – even with the ADC clock at 250kHz.

I have a long way to go before I reach the accuracy levels you see at the geotechnical high end, but I think that’s still good for readings with a humble Arduino ADC!

Addendum 2017-09-24:

Several people have contacted me about their attempts to get this ‘pin-toggling noise’ method working with different Arduinos at higher voltages.  If I had to summarize the kernel of understanding that was missed in the unsuccessful cases it is this:

If you jiggle one part of the system with noise – stabilize the other part.

It does not matter if the noise shows up on aref, or on the sensors output, so long as it is not present in the same form on both.  With the bandgap 1.1v as aref, you can rely on that to be the stable side, so you want the voltage divider with your sensor not to have a capacitor on it, since the sensor side needs to shake by ±2 LSB volts with the pin toggling. The internal reference is slightly different on each individual chip (from 1V to 1.2V), so you’ll need to “calibrate” if you go this route. Don’t forget to throw away the first reading after changing the analog channel, and if you have a high resistance voltage divider, add one ms delay after that first analog read.

If you use the rail voltage as aref (the default) with an un-stabilized voltage divider then your pin toggling current shakes the aref ground in perfect synchrony with the ground line on your sensor, and no matter how many samples you read & decimate you will never get beyond the 10 bit resolution of the ADC. So to use the rail as aref you need a small capacitor across the lower half of your divider so the sensor input to the ADC becomes the stable side: 

Degree Celsius vs. Time with lines offset for easier visual comparison:  The blue line is over-sampled output from a pro-mini clone reading a 100KΩ  NTC Therm/100KΩ series voltage divider. Aref was set to the 3.3v rail, with a 100nF capacitor parallel to the thermistor on the low side.    A 0.12 mA pin-current provided sufficient noise for dithering 1024 readings, delivering an effective resolution of ~0.0028° at 24C. For comparison, the red line is the output from an I2C si7051 sensor on the same logger, with a resolution of 0.01C.

The question of which side should be treated as stable also comes into play when you want to over-sample analog output from more complex sensor circuits. If the sensors supporting electronics are already doing a good job of stabilizing output, say with feedback, caps and some sort of buffer at the end of an amplification cascade, then you have no choice but to set aref to the rail voltage and shake that. I’ve had success with this approach and a complex sensor circuit on a 5v Nano, by pulsing a pin connected to ground through a 12KΩ resistor (~ 0.4 mA of pin current). Of course everything else in your system is feeling this noise to some extent, and this may cause issues with sensitive sensor IC’s, or with micro-controllers other than the 328p.  The effect of the pin current is being limited by capacitance distributed throughout the system, which varies from board to board, so this is definitely a “try it an see” method : when it works it really works, producing smooth curves with no hint of the underlying 10-bit ADC peaking through.

If you see any flat spots or rounded stair steps in your temp. data, especially in areas where the changes are occurring slowly over time, then you know the dithering is not working:  

This is an example of the natural noise problem: the (blue line) thermistor readings achieved high bit depths the refrigerator, but developed flat spots in the room where the changes were happening more slowly. This was a test run with the noise circuit disconnected,which I followed with run using the same code +noise applied so I could compare the two. Doing two runs (with & without dithering) is good general approach to use when testing a circuit that uses oversampling.

Any natural temperature variation over your sampling interval will make it look like your generated noise is sufficient for oversampling, when it is not.  The photo above shows how that this test is almost impossible to do in the refrigerator, because the natural on/off cycle of the compressor generates enough change/time to make oversampling work without dithering. 

With stabilizing capacitors you also have the trickier problem of spotting is the influence of the RC time constant when you only power the voltage divider during readings.  Oversampling before the cap is fully charged will provide more than enough change in the readings to hide inadequate dithering.  In fact, if you scale the capacitor/series resistor combination, and the timing, to sample over the 3T-5T interval, you get good oversampling results with no noise in the system at all.  In some ways, using RC rise time is better than pin toggling when you are using the rail as aref, since it does not fight against the other capacitance distributed around the system to produce a delta on the ADC readings.  I’d use this rather than pin toggling with aref=rail  if it weren’t for the fact that capacitors can have the worst variation coefficients of any electronic component you are ever likely to run into.

Garden variety Y5V ceramics vary by up to 82% over their rated temperature range, and even the X7R’s that most engineers use vary by +/-15%. I could calibrate that thermal variation away, but from an environmental monitoring point of view, the drift over time is a much bigger problem; with caps commonly loosing 10-15% of their rating over the first year (~8900 hours) of operation. There are stable NPO rated ceramic caps out there, but they are only available in relatively small pF sizes, and a good 0.1uF NPO cap will set you back about $7 each even if you buy them in quantity, so that part alone costs more than a decent IC based temperature sensor.

Plastic film capacitors have much better thermal coefficients: Polyphenylene sulfide (PPS ±1.5%) or Polypropylene (CBB or PP ±2.5%)A quick browse around the Bay shows those are often available for less than $1 each, and the aging rate (% change/decade hour) for both of those dielectrics is listed as negligible. The trade off is that they are huge in comparison to ceramics, so you are not going to just sneak one in between the pins on your pro-mini. 

For most rail-as-aref situations, Qwerty’s PWM based dither method (mentioned at the beginning of the post) is a more robust way to dither with cheap ceramic caps, since it can tolerate significant variation in a way that does not affect your accuracy that much – but you still have to keep an eye on the dither circuit settling time. 

Addendum 2017-10-15:

Just came across AN2668 from STMicroelectronics which sums the input signal and trianglar dither signal through an opamp before sending it to the ADC:

Still seems like a lot of work to me, although that apnote does have me wondering if the pin toggle dither noise is actually Gaussian…

The 2016 Cave Pearl Project ‘Year in Review’

That's a chain with 24xDS18b20 sensors pulling only 0.15 mA sleep current. Woot!

That’s a chain with 24 DS18b20 sensors pulling only 0.15 mA sleep current. Woot!

We made great strides in 2016 with development of new calibration procedures and continued refinement of the housings & connectors for multi-sensor builds. At this point I’ve cobbled together more than 130 loggers for the Cave Pearl Project. While I still have a way to go before I reach Gladwell’s 10,000 hours, we haven’t spent $71 million, and I think I’m starting to get the hang of it. Of course, I felt that way last year too, so perhaps that’s just as good as it gets when you are figuring things out as you go along. It’s not unlike that moment when your boss calls you in for the annual performance review, and then starts the meeting by asking you to rewrite your job description, again, because less than half of the work you actually did last year was in the previous one.

Despite all that building, the number of loggers out on deployment leveled off around sixty five as newer units were often used to retire earlier generations out of the fleet. It’s worth noting that those old dogs are all still running, but field logistics have reached the point where we can’t expand the project unless everything on active duty has more than a year of operating life.  Fortunately the latest Pearls are consistently delivering sleep currents around 0.1mA , so builds with more than three AA cells are probably going to run for two years or more. 

No one ever seems to bother our equipment installations...I wonder why that is?

No one ever seems to bother our equipment installations.  I wonder why that is?

The temp. string loggers are currently the most complicated builds, but they have been delivering some very impressive data sets. Despite their flagship status, most are deployed in sites with a nasty mix of salt water and hydrogen sulfide that’s been destroying the marine grade stainless steel anchor weights. As they take a while to assemble, on-site service and redeployment will be the norm for  those logger installations until we get more of them into circulation. Rapid turn-arounds like that were one of the original goals of the project, but that’s only possible in practice because none of our current sensors are affected the bio-fouling that accumulates over a deployment.

There are plenty of new sensor combinations on the todo list as we round out the hydrology tool kit with more cheap & cheerful ‘tattle-tale’ builds. The recent discovery that you can stretch the I2C bus out to 10’s of meters has opened up many new sensor possibilities.  Long-cable MS5803 pressure sensors are now being put into service for borehole logging and if I can reduce the thermal equilibration time, that sensor also shows promise as part of a drop profiler.  From very the beginning, we’ve had our sights set on a cheap reliable CTD with extended logging capability.

Our oldest monitoring stations are reaching the three year mark, which means we are starting to see repeating annual cycles in the data. As an example, here is the temperature and tilt record from one of the reef stations in Akumal Bay:

annualcycles

The occasional hurricane means there are some lovely ( …to a hydrologist  🙂 ) event records peppered throughout, but the real prize is the baseline data that you only get from multi-year deployments. From the builders perspective, those long records also give me a sense of the inter-unit consistency, which is looking  good despite substantial  changes to the loggers over time. As the project matures, more of my time is being dedicated to sensor calibration and normalization.

Though we rarely have enough visibility to capture it with our little point&shoot camera, the underwater procedures are getting smoother too:


Cave Pearl data loggers

At  ~6000 unique IP’s a month, traffic continues to grow, though I will have to manage more than one post per month if the project is ever going to be a real contender for science geek fame.  The UNO logger has now passed the RTC post in the daily rankings, which shows that the number of interested beginners is also increasing.  The US dominates the overall traffic at (with about 50% of the total), and Germany leads the European traffic, often coming in second only to the US.  If other DIY sites are seeing a similar trend, then I think it’s pretty obvious why Germany continues to be the economic powerhouse of Europe

Many years ago I did a short stint as a high school science teacher and some of my good friends are still in the trenches.  Conversations with them often highlight two things:  How their job now depends on standardized test results (so they don’t have much room to change the curriculum), and that resource budgets are shrinking to zero. These are dedicated people who are more than willing to go the extra mile if they can get their hands on the right material.  So I’ve been working to expand the online tutorials into a set that would help a teacher add Arduino based lessons to their curriculum, even if they have do it out of their own pocket:

2016 was another great run of Trish's Instrumentation & field methods course. It's impressive with how quickly the students pick up the Arduino, and everyone got their final projects running though most of them had never held a soldering iron before.

2016 saw another great run of Trish’s Instrumentation & Field Methods course. It’s impressive how quickly the students pick up the Arduino system, and everyone got their final projects running though most of them had never used a soldering iron before.

  1. Build your own Arduino classroom
  2. Arduino UNO Logger for Beginners
  3. Using the UNO for Data Acquisition
  4. ProMini based Data Logger (update)

I’m sure that most people missed the significance of the DAQ post when it came out, but in terms of teaching it’s probably the most important one in the set. The IDE’s new plotting function makes it easy to do live demonstrations in front of a classroom simply by adding one print statement to the code:  nothing I’ve tried before even comes close to this level of simplicity.

I use my  ‘UNO-scope’ all the time now because I can do a quick screen capture  or I can send a series of runs to a serial text monitor. In both cases, calculations can be done on the Arduino before the output is sent, converting the raw ADC readings into something more meaningful like current or power.  Then I can go hunting for unexpected events by graphing it up in Excel.  This approach has let me track down things like SDcard weirdness that are very hard to capture during normal logger operation because they only happen when you pass some threshold inside the flash controller:

Cave Pearl data loggers

I’ve overlaid the bold numbers, but that’s a screen capture of the IDE output. Modern scopes can do this kind of thing too, but you’ll be hard pressed to find one for the price of an Arduino clone on eBay. With a low value shunt resistor, you can push the ADC clock pre-scalars into the 20-40kHz range, which is more than adequate for the kind of diagnostic work I’m doing, and most stuff you’d be trying to demonstrate in a high school physics class. 

DIY Cave Pearl data loggers based on Arduino Microcontrollers

My PVC housings sport smaller separate wells for each sensor as I’m trying to reduce the surface area exposed to pressure & salt water. So far we have only deployed them to ~30m, but deeper sites are on the calendar in 2017.

3D printers are finally reaching a price point where people can afford one to help them pursue other more interesting hobbies, and as key patents continue to expire,  new/old high-end tools are entering the consumer market: Forget 3D-Printed Knick-Knacks.  But it’s worth noting I haven’t used any of these tools on the project because I can still build more durable housings out of plumbing fittings from the local hardware store. Total part cost on those is ~$15, and I probably spend less time making them than I would repairing holes or fixing mesh errors on a constantly evolving 3D printed version. But I’m sure the strength and print quality tipping point will occur eventually… probably when DLP’s with stronger resins reach the kind of price point we are currently seeing for extrusion printers.

I will add more tutorials to that set over time, and hopefully we’ll manage to publish a paper or two this year.  The plan is to put them in open source journals so everyone in the world has access, and if we spin up all the collaborative projects we’ve been planning with other researchers, 2017 promises to be a very busy year.

Addendum 2017-01-10

2016 was also a banner year for the Maker movement in terms of media coverage. So I thought I would add selection of those articles here:

Why the Maker Movement Matters: Part 1, the Tools Revolution

Why the Maker Movement Matters: Part 2, Agility

Sometimes those geeky editorials make me laugh, but even then they are still thought provoking.  It’s also good to see more thoughtful criticism and self reflection going on as the movement matures it’s way through the Hype Cycle (beautifully illustrated by this 100+ year old debate about the Wright brother’s)

Making It

Makerspace: Towards a New Civic Infrastructure

Why I Am Not a Maker

Unfortunately that also means the market has grown to the point that the big boys want a piece of the action. While this probably won’t be another  “Embrace, Extend & Extinguish” situation, commercial players inevitably increase the pressure to commoditize the product into easier to use (& thus more sellable) packages. I can see good arguments to support this but I have some concern about developments (like the ESLOV) which eliminate the user’s exposure to actual code: turning great problem-based learning exercises into plug & play activities.  Unless you let students see ‘under the hood’, they’ll walk away thinking technology is about connecting little black boxes for participation marks.  By now it’s clear that we are heading toward a IoT powered world of self driving carsBaxter bots, and staffless stores.  So I can’t help thinking that unless our young people can handle Arduino level programming tasks, they won’t qualify for a job making toast.

Addendum 2017-04-30

On the topic of media: looks like the one-word journals are finally starting to notice the open source hardware movement. Better late than never, though I suppose with all the mergers going on, the whole makers movement isn’t much more than a rounding error on corporate scale balance sheets.